Multi-channel control methods for switched power converters

ABSTRACT

Cross-channel interference can be a problem in multi-channel switched power systems. A multi-channel switched power converter is provided a reference count generator that generates a reference count signal that is commonly distributed to multiple, pulse width modulating (PWM) channels of the converter. The common reference count signal counts through a sequence of successive timing values. Events in each channel are scheduled relative to the common reference count. Examples of such events include the amplification or de-application of a driving voltage pulse to an inductive means of the channel and the taking of voltage and current measurement samples. Cross channel interference is avoided or minimized by scheduling or re-scheduling events in each active channel so that they do not occur simultaneously in potentially-interfering channels, for example the taking of a current measurement sample in one channel and the turning on of an inductance-driving voltage pulse is a second channel.

FIELD OF THE INVENTION

This invention relates to switching power supplies or converters. Inparticular, this invention relates to a simple, robust switching powersupply which is capable of providing power to a number of differentregulated power sources within a given circuit.

BACKGROUND OF THE INVENTION

Switching power supplies are used to provide power in numerous productssuch as cell phones, camera, PDAs (Personal Digital Assistants),calculators, portable computers and similar types of electronicequipment. Such switching power supplies are quite complex and usenumerous components to provide a number of precisely regulated outputvoltages to power the various integrated circuits and other componentscontained within the product being powered. Relative to the cost and thequality of the products in which they are used, such power supplies areexpensive, bulky and inefficient. Efficiency is important to provide theequipment a long battery life. FIG. 1 shows a typical prior art powersupply used in portable equipment powered by a battery 10. The signalfrom battery 10 is transmitted on lead 10 a to a level translationcircuit 12, which is controlled by a control signal from analog pulsewidth modulated controller 11. The control signal from analog pulsewidth modulator is responsive to the voltage drop across resistor 16 asdetected by signals on conductive leads 17 a and 17 b connecting,respectively, the two terminals of resistor 16 into analog PWMcontroller 11. N-channel MOS transistors 13 a and 13 b are connected tooperate in a complementary fashion. Level translation circuit 12provides a high level voltage to the gate of N-channel transistor 13 ato apply a pulse from battery 10 to one input terminal of coil 15. Theother input terminal of coil 15 is connected to one terminal of resistor16. The other terminal of resistor 16 is connected to load capacitor 18,which contains a charge at the voltage necessary to supply theparticular circuitry being powered by this portion of the power supply.The analog PWM controller 11 measures the current through resistor 16and controls the ON time of N-channel MOS transistor 13 a. N-channel MOStransistor 13 b is driven by the complement of the signal driving thegate of N-channel MOS transistor 13 a and turns on to pull the inputlead of coil 15 to ground and to shut off the current required to besupplied through resistor 16 to the power supply. Internal circuitry ofanalog pulse width controller 11 is shown schematically in FIG. 2.

As shown in FIG. 2, current source 20 provides a charging current tocapacitor 21 to generate a ramp voltage across this capacitor. This rampvoltage is provided to the positive input lead of differential amplifier22 a, the negative input lead of which receives the output signal fromdifferential amplifier 22 b. The positive input lead of amplifier 22 bis connected to the load capacitor 18 and carries a signal representingthe voltage across the load capacitor 18. The negative input lead ofdifferential amplifier 22 b is connected to the node between resistors23 a and 23 b making up a voltage divider (one terminal of which isconnected to a reference voltage VRef and the other terminal of which isconnected to the output lead of differential amplifier 22 b). Thus whenthe output voltage across capacitor 18 is less than the voltage at nodeA between resistor 23 a and resistor 23 b, the output voltage fromdifferential amplifier 22 b goes to a low level. This low level outputvoltage is provided to the negative input lead of amplifier 22 a,causing amplifier 22 a to produce a positive output pulse. This positiveoutput pulse is transferred to coil 15 to provide a charging current tocapacitor 18. With time, the charge on capacitor 18 increases until thevoltage across capacitor 18 exceeds the voltage on node A. At this pointthe output voltage from differential amplifier 22 b goes to a highlevel, so that the voltage at the negative input lead of differentialamplifier 22 a exceeds the voltage on the positive input lead ofdifferential amplifier 22 a, causing the output voltage from amplifier22 a to go a low level, and thus preventing further charging ofcapacitor 18. The voltage across coil 15 is negative, reflecting thenegative rate of change in current in response to the trailing edge ofthe pulse from amplifier 22 a going from a high level to a low level.The current through coil 15 does not change instantaneously due to themagnetic field of the coil but rather gradually changes with time. Thistype of power supply, which is characterized by a current source drivinga capacitor, is known as an analog buck converter. Each MOSFETmodulation cycle is formed by the precision comparator and the erroramplifier. Such a power supply is difficult to scale and integrate intoan integrated circuit and is typically fabricated using dedicated analogprocess technologies at captive semiconductor foundries.

Accordingly, what is needed is a power supply which provides differentlevel precision voltages and at the same time and is simple to implementwith a smaller number of components than in the prior art. Such a powersupply must also be relatively inexpensive, robust and reliable.

SUMMARY OF THE INVENTION

A multi-channel switched power converter in accordance with theinvention has a reference count generator that generates a referencecount signal that is commonly distributed to multiple, pulse withmodulating (PWM) channels of the converter. The common reference countsignal counts through a sequence of successive timing values. Events ineach channel are scheduled relative to the common reference count.Examples of such events include the application or de-application of adriving voltage pulse to an inductive means of the channel and thetaking of voltage and current measurement samples. Cross channelinterference is avoided or minimized by scheduling or re-schedulingevents in each active channel so that they do not occur simultaneouslyin potentially-interfering channels, for example the taking of a currentmeasurement sample in one channel and the turning of aninductance-driving voltage pulse in a second channel.

Other aspects of the invention will become apparent from the belowdetailed description.

This invention will be more fully understood in conjunction with thedrawings taken together with the following detailed description.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows the basic construction of a prior art analog buckconverter.

FIG. 2 shows the details of the analog PWM controller 11 in FIG. 1.

FIG. 3 shows the basic structure of a pulse width modulation controllerutilizing a ring oscillator in accordance with this invention.

FIG. 4 shows an alternative implementation of a pulse width modulatedcontroller in accordance with this invention.

FIG. 5 shows an example of waveforms of the type generated using thestructure of FIG. 3.

FIG. 6 shows a ring oscillator used in accordance with this inventiontogether with a switching matrix used to select the particular outputsignals from the ring oscillator to be provided to the two input leadsof an exclusive OR gate 63 to generate a pulse width modulated signal.

FIG. 7 shows the waveform of signals generated using the structure ofFIG. 6.

FIG. 8 shows the waveforms generated using the structure of FIG. 6 withan inverter connected between the output lead from each even-numberedinverter in the ring oscillator and the pass-transistors driven by thesignals G, H, I, J and K.

FIG. 9 shows the relative delay times obtained when signals are selectedfrom different combinations of pairs of inverters in the ring oscillatorof FIG. 6.

FIG. 10 shows one circuit for controlling the selection of theparticular gates to transfer a selected pair of signals to the exclusiveOR gate 63 in FIG. 6 to generate a pulse width modulated signal.

FIG. 11 shows another circuit used to generate the pulse width modulatedsignal from exclusive OR gate 63 in FIG. 6 in accordance with theprinciples of this invention

FIG. 12 is a block diagram of switching power supply controller 1200, inaccordance with one embodiment of the present invention.

FIG. 13 is a block diagram showing interface signals of digital pulseconverter wrapper 1201, according to one implementation.

FIG. 14 is a block diagram showing interface signals of analog todigital converter 1206, according to one implementation.

FIG. 15 is a block diagram showing the interface signals of Kelvintemperature sensor (KTS) 1500.

FIG. 15A is an exemplary circuit for KTS 1500 in accordance with oneimplementation.

FIG. 16 is a timing diagram illustrating the quad-slope (i.e., dualconversion) analog to digital conversion (ADC) operations carried out inthe QSADC module 1211 a, which is contained within touch screeninterface 1211.

FIG. 17 is a block diagram showing the interface signals of QSADC module1211 a in one implementation.

FIG. 17A is a block diagram showing the interface signals of QSADC 1211a with touch screen interface 1211 in one implementation.

FIG. 18 is a top-level block diagram of QSADC module 1211 a, includinganalog block ANLG 1801, control block CNTRL 1802 and DOWN/UP COUNTERblock 1803, according to one embodiment of the present invention.

FIG. 18A shows one implementation of analog block 1801 of QSAD module1211 a shown in FIG. 18.

FIG. 18B shows one arrangement of a four-contact touch-screenapplication.

FIG. 18C shows one arrangement of a five-contact touch-screenapplication.

FIG. 19 is a block diagram summarizing the modules in switching powersupply controller 1200, providing a control loop for a battery or powersupply management application.

FIG. 20 illustrates regulation of output voltage Vout using inductorcurrent IL and sensing output voltage Vout, according to one embodimentof the present invention.

FIG. 21 illustrates the approximation of a weighted average inductorcurrent by duty cycle.

FIG. 22 illustrates a method under the present invention for estimatingthe parasitic resistance of an MOS switch.

FIG. 23 shows flow diagram 2300, illustrating a control method accordingto one embodiment of the present invention.

FIG. 24 illustrates a low-frequency closed loop and high-frequency openloop control method, according to one embodiment of the presentinvention.

FIG. 25 is a diagram showing switching power supply controller 1200being used in a battery and power supply management application in, forexample, a personal digital assistant (PDA).

FIG. 26 illustrates the operation of a control loop in accordance of thepresent invention.

FIG. 27 illustrates the low-frequency closed loop and high-frequencyopen loop control method of FIG. 24, showing the signature input valuesof controlled variables and an open-loop response, according to oneembodiment of the present invention.

FIG. 28 illustrates a sequential transient recovery control method, inaccordance with one embodiment of the present invention.

FIG. 29 illustrates a buck converter.

FIG. 30 through FIG. 33 show curves of current with respect to time fora supply circuit such as circuit 49, based on the application ofdifferent gate drive voltages to the transistors.

FIG. 34 is a block diagram showing the interface signals of clockgenerator 12223.

FIG. 35 shows the port table for the clock generator block shown in FIG.34 and FIG. 12.

FIG. 36 shows an exemplary pulse width modulation implementation, whichincludes a pulse width generator and a sequencer, for digital pulseconverter 1201.

FIG. 36A is an exemplary implementation of the pulse width generator ofFIG. 36.

FIG. 36B is an exemplary implementation of the sequencer of FIG. 36.

FIG. 36C shows an exemplary feedback control system.

FIG. 36D shows another exemplary feedback control system.

FIG. 37 shows a timing diagram for one digital to pulse converter framefor DPC 1201 in accordance with one implementation.

FIG. 37A shows exemplary interface signals for digital pulse converter1201 for one implementation.

FIG. 37B illustrates an exemplary circuit implementation for DPC 1201,which includes a CAM.

FIG. 37C illustrates one exemplary implementation for the CAM of FIG.37B.

FIG. 37D illustrates an exemplary circuit implementation for outputlogic of the CAM of FIG. 37C.

FIG. 37E illustrates an exemplary timing diagram for the CAM of FIG.37C.

FIG. 38 shows an exemplary implementation of a Grey counter.

FIG. 38A shows another exemplary implementation of a Grey counter.

FIG. 38B shows an exemplary implementation for a flip flop of FIG. 38 orFIG. 38A.

FIG. 38C shows an exemplary implementation for another flip flop of FIG.38 or FIG. 38A.

FIG. 38D shows an exemplary circuit implementation for a logic gate.

FIG. 38E shows an exemplary circuit implementation for another logicgate.

FIG. 38F shows an exemplary circuit implementation for a multiplexer.

FIG. 38G shows an exemplary circuit implementation for binary to Greyand Grey to binary conversion.

FIG. 38H illustrates an exemplary implementation for a digital pulseconverter.

FIG. 38I illustrates another exemplary implementation for a digitalpulse converter.

FIG. 39 shows a plot of typical gate drive waveforms which may beapplied to, for example, the gates of the transistors of circuit 49 andthe resulting voltage at terminal S.

FIG. 40 is a block diagram of SHM 1207 in one implementation.

FIG. 40A is a functional schematic illustrating voltage and currentsampling for one implementation.

FIG. 40B is a circuit schematic illustrating voltage and currentselection for one implementation.

FIG. 40C is an exemplary interface signal block for SHM 1207 inaccordance with another implementation.

FIG. 40D is a block diagram of SHM 1207 in another implementation.

FIG. 40E is a functional schematic illustrating voltage and currentsampling for another implementation.

FIG. 40F is a circuit schematic illustrating voltage and currentselection for another implementation.

FIG. 40G is a clock generation circuit in accordance with anotherimplementation.

FIG. 40H is a voltage divider in accordance with another implementation.

FIG. 40I is a voltage multiplier in accordance with anotherimplementation.

FIG. 40J is an exemplary interface signal block for an I/O circuit inaccordance with another implementation.

FIG. 40K is a multiplexer scheme in accordance with anotherimplementation.

FIG. 41 is a block diagram of the regulation control module (REG) ofFIG. 12 according to one embodiment of the invention.

FIG. 42 shows a gate drive waveform with respect to time.

FIG. 42A shows a boost converter circuit.

FIG. 42B shows two switching waveform in time off-set relationship.

FIG. 43 shows waveform A, B and C illustrating the current which isoutput for three gate drive scenarios of a switching power supply.

FIG. 43A shows a plot of the voltage with respect to time at a terminalS intermediate the upper and lower transistors in buck converter 49 ofFIG. 29.

FIG. 43B and FIG. 43C show plots of the voltage at terminal S for twodifferent duty cycles of FET 50 in circuit 49.

FIG. 44 shows a circuit for generating a high voltage to drive a coldcathode fluorescent light bulb.

FIG. 44A shows two exemplary sets of gate drive waveforms of the typewhich could be applied to the gates of the transistors of circuit1.2.2.12.

FIG. 45 shows a buck converter circuit.

FIG. 45A shows gate drive waveforms for the transistors in FIG. 45 andcorresponding current and output voltage waveforms.

FIG. 45B shows a boost circuit.

FIG. 45C shows gate drive waveforms for the circuit of FIG. 45B andcorresponding current and output voltage waveforms.

FIG. 46 shows switching power supply controller 1200 connected toregulate the operation of two switching power supplies.

FIG. 46A shows a plot of current with respect to time for one cycle of aswitching power supply.

FIG. 46B shows a buck power supply circuit.

FIG. 46C shows a plot of current with respect to time for one cycle of aswitching power supply circuit.

FIG. 46D shows a plot of current with respect to time for one cycle of aswitching power supply.

FIG. 46E shows a plot of current with respect to time for a switchingpower supply circuit.

FIG. 47 shows switching power supply controller 1200 connected to aplurality of power supplies.

FIG. 48 shows in block diagram form a processor coupled to two switchingsupplies, a nonvolatile memory and a Kelvin temperature sensor, with abattery connected to one of the switching power supplies.

FIG. 48A shows a curve of battery capacity with respect to temperaturefor a typical battery.

FIG. 49 shows a block diagram of a power supply system for use inconjunction with a cellular telephone.

FIG. 49A shows a block diagram for a solar cell array coupled to a powersupply being controlled by switching power supply controller 1200 of thepresent invention.

FIG. 50 is a block diagram of a spreader divider unit 2482.4 accordingto one embodiment of the invention.

FIG. 50A is a circuit diagram of a spectral spreader 210.1 for thespreader divider unit 2482.4 of FIG. 50 according to one embodiment ofthe invention.

FIG. 51 is a block diagram showing the interface signals of QSADC module1211 b in another implementation.

FIG. 51A is a top-level block diagram of QSADC module 1211 b, includinganalog block 2001.4, control block 2002.4, and up/down counter block2003.4, according to another embodiment of the present invention.

FIG. 51B shows one implementation of analog block 2001.4 of FIG. 51A.

FIG. 51C illustrates a block diagram showing exemplary interface signalsbetween touch screen interface 1211 and the other blocks of switchingpower supply controller 1200.

FIG. 51D shows a flowchart for performing diagnostics of QSADC module1211 a.

FIG. 51E shows a functional block diagram corresponding to theimplementation of FIG. 51B.

FIG. 51F illustrates the circuit connections of analog block 1801 for aninitial measurement state.

FIG. 51G illustrates the circuit connections of analog block 1801 tointegrate the voltage due to contact with the Y coordinate sheet in afour-contact implementation.

FIG. 51H illustrates the circuit connections of analog block 1801 todigitally convert the voltage due to contact with the Y coordinate sheetin a four-contact implementation.

FIG. 51I illustrates the circuit connections of analog block 1801 tointegrate the voltage due to contact with the X coordinate sheet in afour-contact implementation.

FIG. 51J illustrates the circuit connections of analog block 1801 todigitally convert the voltage due to contact with the X coordinate sheetin a four-contact implementation.

FIG. 51K illustrates the circuit connections of analog block 1801 tointegrate the voltage due to contact with the X-Y coordinate sheet in afive-contact implementation due to horizontal position.

FIG. 51L illustrates the circuit connections of analog block 1801 todigitally convert the voltage due to contact with the X-Y coordinatesheet in a five-contact implementation due to vertical position.

FIG. 51M illustrates the circuit connections of analog block 1801 tointegrate the voltage due to contact with the X-Y coordinate sheet in afive-contact implementation due to horizontal position.

FIG. 51N illustrates the circuit connections of analog block 1801 todigitally convert the voltage due to contact with the X-Y coordinatesheet in a five-contact implementation due to horizontal position.

FIG. 51O illustrates the circuit connections of analog block 1801 todetect continuity between the resistive sheets in a four-contactimplementation.

FIG. 51P illustrates the detection of contact for a five-contactimplementation.

FIG. 52 illustrates a block diagram of LED control block 1214.

FIG. 53 is a block diagram for the watchdog module of FIG. 12 accordingto one embodiment of the invention.

FIG. 54 is an exemplary functional diagram of Internal Voltage Supply(IVS) 1209 in accordance with one implementation.

FIG. 54A is an exemplary interface block diagram for IVS 1209 inaccordance with another implementation.

FIG. 54B is another exemplary functional diagram of IVS 1209 inaccordance with another implementation.

FIG. 54C is an exemplary flowchart for a power-up sequence for IVS 1209in accordance with another implementation.

FIG. 55 is an exemplary functional diagram of NFET driver module 1202 inaccordance with one implementation.

FIG. 55A illustrates exemplary interface signals of NFET driver module1202 in accordance with another implementation.

FIG. 55B illustrates exemplary interface signals of NFET driver module1202 in accordance with another implementation.

FIG. 55C shows an exemplary functional diagram for the implementation ofFIG. 55B.

FIG. 55D illustrates an application employing internal buffers todirectly drive an external coil for the implementation of FIG. 55B.

FIG. 55E illustrates an application employing internal buffers to driveexternal FETs for the implementation of FIG. 55B.

FIG. 55F shows an exemplary on-chip configuration block diagram for theimplementation of FIG. 55B.

FIG. 56 is a block diagram of central processing module (SYS) 1205 ofFIG. 12 according to one embodiment of the invention.

FIG. 57 shows a circuit model of converter 2570 of FIG. 25.

FIG. 58 is a flow chart illustrating the steps of control algorithm5800, in accordance to with one embodiment of the present invention.

FIG. 59 illustrates the operation of control algorithm 5800 of FIG. 58.

DETAILED DESCRIPTION OF THE EMBODIMENT(S)

The following detailed description is intended to illustrate theembodiments discussed herein and is not intended to be limiting of thescope of the present invention. Other embodiments of this invention willbe apparent to those skilled in the art in view of this disclosure.

Section 1.0 Overview of the Architecture of a Converter (DigitallyTrimmed Multi-stage SPS) Having a Synchronous Sampling Multiple-outputController, Functional Description of Each Block

The present invention is applicable to a power converter and powermanagement peripheral that integrates a set of power-management relatedfunctions. Switching power supply controller 1200, shown in FIG. 12 inblock diagram form, illustrates an implementation of a switching powerconverter controller product, according to one embodiment of the presentinvention. The system implements substantially all of a powerconverter's expected functions, such as maintaining a steady outputvoltage (constant voltage power supply) that is substantiallyindependent of the current drawn from the power supply or maintaining asteady output current (constant current power supply) that issubstantially independent on the load applied to the power supply,deciding when to shed loads, and measuring the state of charge in abattery, charging the battery, and performing battery sequencing. Inaddition, the system performs a number of other peripheral managementfunctions, such as digitizing a touch panel, scanning a keyboard, andconditioning reset signals received from the rest of the system. In oneembodiment, watch dog timer functions are provided to allowpower-cycling and to provide reset signals to different subsystems ofthe product, in response to an exception condition, such as a softwarelock up or even a hardware lock up (e.g., an SCR latch up or adisruption due to an IO transient). The system of the present inventionalso controls LEDs that indicates the statuses of various subsystems.Thus, a system according to the present invention can handle a largenumber of functions that a portable electronic device (e.g., a PDA)performs when the processor of that PDA is powered down, such as betweenpen strokes (while the user is writing a note on the PDA), or when thePDA is displaying information, but no input or output is expected. Inthis implementation, the product includes a digital control subsystemthat controls various digital interfaces, including reset signals, watchdog timers, enable signals, status signals for indicating individualstatus of internal power supplies. Communication among elements of thesystem is provided through a communication interface. In one embodiment,the present invention includes a time base generation circuit and adigitizer interface to a keyboard.

Referring to FIG. 12, substantially all of the digital interfacefunctions are performed by an internal 8051 or a comparablemicroprocessor which is included in central processing module (SYS) 1205and which communicates with the outside system with a number ofinterfaces through a number f general purpose input/output (GPIO)signals that can be defined in software in the microprocessor and mappedflexibly to individual input or output pins or terminals of theintegrated circuit. For example, each pin can be defined to carry a hightrue or a low true signal, and can be defined to be an input signal oran output signal. In addition, each pin can be assigned to any of theindividual power supply elements and can be shared so that more than onepower supply can receive the same control signal from a single pin. Inthis manner, for example, a single enable signal may be shared amongstmultiple power supplies. The GPIO signals can be used to implement inputand output signals of some of the peripheral devices. For example, thekeyboard scanner functions interact with an external keyboard throughthe GPIO signal interface. The watch-dog timer may receive input signalsand status information from circuits external to the integrated circuit.Also, interrupt signals can share pins with the GPIO signals. Thisimplementation thus provides a rich set of functionality to allow theproduct to be capable of being made “plug-compatible” with prior artsolutions that were based on separate discrete devices. Centralprocessing module (SYS) 1205 handles serial communication on and off thechip. Every signal which can be implemented as a GPIO signal, except aninterrupt signal, can also be implemented as a serial communicationsignal. In an application designed for prior art devices, serialcommunication can be used to access a keyboard and coulombmetricmeasurement capabilities. The GPIO signals can be used to turn on andoff individual power supplies and to monitor their statuses.

A computation circuit, which may be provided as a very long instructionword processor (“VLIW engine”) to simplify decoding logic, is includedin regulation control module (REG) 1204 (labelled in FIG. 12 as “REG”).The VLIW engine executes low level commands from central processingmodule (SYS) 1205 (labeled in FIG. 12 as “SYS”). Regulation controlmodule (REG) 1204 receives and interprets the measurements of voltagesand currents from the various circuits supplied by the regulated powersupply pins. Using a number of different types of control loops,regulation control module (REG) 1204 provides to digital pulse converter(DPC) 1201 commands in the form of precisely modulated signals, whichthen are used to operate external components through NFET driver module1202. Together with the external components, switching power supplycontroller 1200 forms one or more power converters of various designatedtopologies.

One unique feature of the system according to the present invention isthat the system is programmable. In addition to allowing every function,mode and regulation parameters to be preset, the specifications ofexternal components can be stored internally in the integrated circuitand used to perform the regulator functions. Thus, regulation controlmodule (REG) 1204 operates with a large amount of information about thecharacteristics of the power converter components external to switchingpower supply controller 1200. Unlike a prior art switching power supplycontroller circuit implemented in analog technology, a power supplycontroller circuit of the present invention “knows” the output voltageit's trying to attain (“target voltage”) and the error (i.e., thedifference between the current output voltage and the target voltage).Typically, prior art analog converters correct the error usingalgorithms that are independent of the current switching duty cycle orthe value of the input voltage. Such prior art converters also do notinternally store information about the external components. An engineerattempting to design with such a prior art power supply controllercircuit can only provide “hints” to the power supply controller circuitin the form of compensation networks. In contrast, a power supplycontroller circuit according to the present invention has manyadvantages and benefits over the prior. For example, having informationregarding the current output voltage and duty cycle, the target voltageand the internally stored parameter values of the external circuitallows the power supply controller to calculate the potential dutycycles that would result in the correct output voltage to high precisionand to select a realistic response that observes the constraints of theexternal circuit (e.g., the saturation current of an inductor). Properresponse to external circuit anomaly limits the amount of noise that'sreflected to the power supply controller circuit. For example, bycontrolling the transient noise currents through the power controllercircuit, a battery or a long wire that provide the supply voltage to thepower controller circuit could experience and possibly radiate lesselectromagnetic interference (EMI).

A programmable keyboard scan function is implemented in centralprocessing module (SYS) 1205, and allows individual keys to be scannedor detected after a key has been pressed to awake the keyboard scanningcircuit from a stand-by state. A watch dog function is performed withsoftware in central processing module (SYS) 1205, and providesfull-featured watch-dog timer functions. Watch dog timer functions maybe used to handle software faults in a system. For example an externalprocessor running a complex operating system (e.g., Win CE) mayperiodically briefly assert (“toggle”) a signal on a specified pin. Thewatch dog timer resets its timer whenever the signal on the specifiedpin is toggled. When a malfunction occurs in the software of the hostmicroprocessor, such that the signal on the specified pin is not toggledwithin the scheduled time, then a set of pre-programmed actions aretaken to recover operation of the external system processor. Theseactions range from simply resetting the processor or another circuitelement, or cycling the processor's power off and on. Power-cyclingremoves current from parasitic SCRs that may be present in the processoror other integrated circuits, thereby affecting a recovery from asemiconductor latch-up state. Other anomalies that would normally not berecoverable using logic or reset signals may also be corrected throughpower cycling.

Central processing module (SYS) 1205 controls external status LEDs, or asingle multi-color LED. Internal reset logic which is included ininternal voltage supply 1209, provides a power-on reset to allowinternal clocks and internally generated voltages to stabilize prior tooperation. This is a different and distinct concept from the host resetconditioning feature described below, which is implemented in softwareand runs on the internal microcontroller. The host reset conditioningsoftware conditions the external system's reset signal with the statussignals of the power supplies, and external signals at the product'sinput pins.

Touch screen interface 1211 uses a dual-slope technique to read the Xand Y coordinates of a resistive touch panel display. In a PDA, thisdigitization operation, because of the large area of the display, isquite subject to noise induced by the back light. In the prior art, atypical back light is implemented by cold cathode fluorescent light(CCFL), which is essentially a fluorescent light bulb that does not havea cathode heater to raise the energy of the internal gas for ionizationat a low voltage. In CCFL, for a typical device, initial ionization(“ignition”) is brought about using an AC voltage of 700 volts or more,and an AC voltage of over 300 volts is used thereafter to maintain theignition. The high voltage AC waveform that is driving the CCFL is apotentially serious source of noise to the touch panel, which is made upof panels of resistive material and is located only a few millimetersaway directly in front of the display. Touch screen interface 1211 usesa quad-slope analog-to-digital converter circuit which operatessynchronously with the back light voltages generated by the power supplycontroller. By operating synchronously with the back light, the noisefrom the back light is integrated over an even number of cycles, therebyeffectively removing it without a of complex filtering or an algorithmicapproach.

Internal voltage supply 1209 is the power supply for switching powersupply controller 1200, deriving from either one of two external supplypins, or one of two external battery pins the internal voltages requiredfor the operations of the various blocks, including central processingmodule (SYS) 1205 and regulation control module (REG) 1204. In additionto providing the various voltages required for the various subsystems,internal voltage supply 1209 also provides the crystal oscillatorfunction (other than the crystal, which is off-chip) various chargepumps for creating internal supplies, and comparators to indicate whenthe supplies are stable for use by internal processing elements.

Sample and hold module 1207 includes an array of sample and holdcircuits and scaling circuits. Sample and hold module 1207 monitors thevarious points within the output sections of the power supplies,measuring voltages and currents, input voltages, and temperatures atvarious points. Sample and hold module 1207 provides its data, onesample at a time, to analog to digital converter 1206 (a system sharedresource), which converts the external analog samples into digitalsamples. The digital samples are used by regulation control module (REG)1204, or passed upstream to central processing module (SYS) 1205. Inthis embodiment of the present invention, the analog to digitalconverter and the sample and hold structures are based on either ratiosof capacitors or unity gain. Thus, calibration can be achieved using asingle calibration.

NFET drivers module 1202 include a set of output drivers which operatein one of two modes. The first mode is to drive external power MOSFETdevices. In this mode, each output voltage is associated with two setsof driver circuits; one for driving the control FET and the other onefor driving the sync FET. For modest currents, the two driver circuitsmay be used in another mode of operation where they are driven together,and their output signals may be connected together, to drive an externalcoil directly in a power switching function.

A power supply utilizing the present invention can be completelyprogrammable, i.e., no external discrete components have to be chosen toset voltages and currents and a single chip can perform many differentfunctions across many different products. Such programmability has thebenefit of reducing the component count on a circuit board. It also hasthe benefit of reducing the number of parts that an individualmanufacturer of an end product would have to stock since this samedevice can be used across many different applications. All of thesefeatures can be preset, either at the time the chip is manufactured, atthe time that it is delivered, i.e., through a distributor, usingprogramming techniques similar to those used with programmable logicarray devices. Alternatively, the system of the present invention can beprogrammed by a customer even on the circuit board during the in-circuittest phase or final test for their product. Further, even though allthese functions can be programmed, the values programmed duringmanufacturing and test are used merely as initial conditions and may bechanged dynamically (i.e., during operation) by the system containingthe power supply controller. For example, the system can reprogram theinternal voltages and currents dynamically, as is often required incomplex microprocessors which require different voltages for differentmodes of operation. Reprogramming activity can also be done for thepurposes of dimming displays and controlling motors. Digital-to-analogconversion functions can be implemented by rapidly changing the outputvoltages of the regulators. Switching power supply controller 1200features a completely programmable start-up and shut-down sequence, soas to allow a system using power supply controller 1200 to be started ina sequence that would avoid a latch-up conditions. For example, the I/Ostructure of a microprocessor is often required be powered before themicroprocessor core. To reverse the order could result in damage ordestruction of the microprocessor or cause a malfunction. A specificpower sequence is also required for powering down these devices. Inprior art solutions, this timing is set rather crudely by resistors andcapacitors or not at all.

The individual power supplies controlled by switching power supplycontroller 1200 can be programmed for a wide variety of topologies, sothat if a voltage which is higher than the input source, lower than theinput source, or sometimes the same as the input source can beaccommodated. For example, a boost converter topology is used for apower supply having an output voltage that is higher than the inputvoltage, a buck converter topology is used for a power supply having anoutput voltage that is lower case than the input voltage, and a sepictopology is used for a power supply having the same input and outputvoltages. When very high voltages are required (i.e., a cold cathodefluorescent light bulb or even a photographic strobe in a digitalcamera), topologies such as half-bridge may be used. All of these can beprogrammed and any number of them can be present in the design at anytime, and any combination of these topologies may be supportedsimultaneously.

Switching power supply controller 1200 of the present invention can alsoperform dimming using pulse width modulation. This ability is importantfor cold cathode fluorescent lights because, typically, a simplereduction of current provides insufficient energy to ionize the entiredisplay, resulting in the so called “thermometer effect” where only partof the back light is actually illuminated. Another application where PWMdimming is desirable is in white LEDs. White LEDs suffer an estheticallyunacceptable shift in hue or color, as a function of current. Usingpulse-width modulation for dimming white LEDs maintains a constantcurrent during “on” time, so that a constant color is maintained over alarge dimming range. Switching power supply controller 1200 also hasinput pins for use in temperature compensation: an internal temperaturesensor is included, as well as external pins for reading an externaltemperature sensor. Temperature compensation is used in battery-chargingto tailor a rate of charge, to respond to unsafe environmentalconditions, to detect fault conditions and to prevent the destruction ofexternal batteries or damage due to excessive heat build-up. Theinternal battery charger algorithms accommodate a number of differentchemistries (e.g., lithium ion). Any chemistry can be accommodated sincebattery charging algorithms are provided in software to be executed incentral processing module (SYS) 1205.

Switching power supply controller 1200 also allows selection betweendifferent batteries as its power supply. Central processing module (SYS)1205 can be programmed to use an external battery first, thus preservingits internal battery for emergency situations or while the externalbattery is being changed. It can also automatically choose to charge theinternal battery first, and then external accessory batteries second.Central processing module (SYS) 1205 computes how much energy isavailable and charge both batteries at the same time or use them inparallel. Another function provided by a product using the presentinvention is the ability to provide the voltage, current andcoulombmetric data to the outside system. This permits independentdirect control over the voltage regulator functions and intermediatereadings of the amount of energy available in a battery before anindividual threshold. A system encompassing the present invention alsomaintains a charge acceptance history. This is useful in determiningwhether the battery may be fully charged. It's also an early indicatorof battery wear out and provides a basis for limiting overcharge eventswhere the battery may be indefinitely charged because of some sort ofmalfunction.

In accordance with one aspect of the present invention, the individualswitching waveforms of the product are carefully staggered so that theamount of energy that is derived from an external power source, forexample, a battery, is made as uniform as possible around all of thedifferent outputs that require power. This effectively raises thefrequency of the current required from the external source and mayreduce the peak current demanded from an external source, which reducesthe amount of noise on that external source and also reduces the noiseradiated from the interconnecting wires to the external source and makesthe noise easier to filter. Additionally, a spread spectrum approach isapplied to internal frequencies. This reduces the net energy at a givenfrequency from external switching power supply functions. That is to saythat instead of always operating at a constant frequency, the spreadspectrum feature allows the switching frequency to be varied rapidly andthat frequency modulated using industry standard patterns so that theenergy at any particular frequency is reduced. The spread spectrumfeature effectively reduces noise that would be experienced by anassociated radio either within a product using the present invention,within the end product that this chip is implemented within or otherproducts that are nearby. The frequency of operation can also bedetermined by external sources. In one embodiment, a 32 kilohertzcrystal oscillator generates all the internal clocks and provides a 32kilohertz time based output for use by other elements within the system.According to another feature, the system provides an external clockwhich is guaranteed to be 20 nanoseconds away from the nearest switchingedge. This unique feature allows the external system to conduct samplingsynchronously with the switching power supply functions in the same waythe system synchronously samples to reduce switching noise internally.Normal switching prior art switching power supplies, since they have noknowledge of what duty cycle they're going to provide in any givencycle, have no way to provide this information ahead of time. Since allof the pulse width modulators used in the present invention are digital,this is achieved by another control signal that is scheduled within thearray of control signals that make up all the pulse width modulators.

Referring to FIG. 12, clock generator macro 1223 (CLKGEN) generatesclock signals required by the modules illustrated in FIG. 12. The clockgenerator module 1223 is illustrated in greater detail in the high levelblock diagram of FIG. 34 which shows the interface signals. The clockgenerator generates the clocks based on a source clock signals providedby digital pulse control module 1201. These source clock pulses arereceived by clock generator module 1223 over CST[9:0] output bus 1223.2and the output ports PLOCK and PLLCK from DPC 1201 which are provided oninterfaces 1223.4 and 1223.1, respectively. In the table below, thefrequencies of the signals on the CST bus 1223.2 are illustrated.

Referring to the various interfaces to clock generator 1223, the CSTinterface from DPC module 1201 is comprised of a 10-bit bus indicated byreference character 1223.2 and FIG. 34. The bus is connected to theoutput of the GREY counter in DPC 1201 and provides the majority of thesource clocks for clock generator 1223. Table 102A shows the frequencyof the signals on each of the bus lines CST[0] to CST[9].

TABLE 102A CST Bit Frequency khz CST[0] 134,217.728 CST[1] 67,108.864CST[2] 33,554.432 CST[3] 16,777.216 CST[4] 8,388.608 CST[5] 4,194.304CST[6] 2,097.152 CST[7] 1,048.576 CST[8] 524.288 CST[9] 524.288

The 2-bit bus PD_OUT[1:0] indicated by reference character 1223.3controls the mode of operation of the generated clock signals. Powermodes of operation are designated as Standard, Low Power and Shut Down.Table 102B below shows the modes as function of the signals on the 2-bitbus.

TABLE 102B PD_OUT[1] PD_OUT[0] Mode 0 0 Shut Down 0 1 TBD 1 0 Low Power1 1 Standard

The clock signal to analog to digital converter 1206 is provided overline 1223.7 and the frequencies at the various modes are indicated asset forth below in Table 102C. The Low Power and Standard modefrequencies are derived from the 538,870.921 khz DPLL clock through adivision by 5.

TABLE 102C Mode Frequency (khz) Shut Down 0 Low Power 107,374.1824Standard 107,374.1824

The clock signal to the sample and hold block 1207 is provided over line1223.9. The frequency which is needed for the various modes of operationis shown in Table 102D below. The Standard mode frequency is derivedfrom bit CST[4] bit, and the Low Power mode frequency is derived from adivision of the CST[8] bit by 2.

TABLE 102D Mode Frequency (khz) Shut Down 0 Low Power 262.144 Standard8,388.608

The clock signal to the internal voltage supply 1209 is provided overline 1223.8 and the frequencies needed for the various modes ofoperation are illustrated in Table 102E below. The Standard modefrequency is derived from the CST[4] that, in the Low Power modefrequency is derived by the division of the CST[8] bit by 2.

TABLE 102E Mode Frequency (khz) Shut Down 0 Low Power 1,048.576 Standard0

The clock signals for touch screen interface 1211 are provided by theQSADC_CLK. The frequencies for the various modes of operations are shownin Table 102F below. The Low Power and Standard mode frequencies arederived from the CST[8] bit through a division by 2.

TABLE 102F Mode Frequency (khz) Shut Down 0 Low Power 262,144 Standard262,144

Regulation control module (REG) 1204 is provided clock signals for theStandard, Low Power and Shut Down modes by the clock generator 1223, andthe frequency for each is indicated in Table 102G below. The frequenciesfor the various modes are derived, for example the Standard mode byusing CST[0] bit and the low power operating frequency is derived frombit CST[8] through a division by 4. In addition, CLKGEN 1223 providesthe SHM CLK and SYS CLK clock signals to regulation control module (REG)1204.

TABLE 102G Mode Frequency (khz) Shut Down 0 Low Power 131.072 Standard134,217.728

Table 102H illustrates the modes and frequencies for those modes for theclock signal provided to central processing module (SYS) 1205. Asillustrated in Table 102H, in the Shut Down mode the frequency is 0. Forthe Low Power and Standard modes, the frequencies are identical and theyare derived from CST[2].

TABLE 102H Mode Frequency (khz) Shut Down 0 Low Power 33,554.432Standard 33,554.432

The LED_CLK is a clock used for the LED block which is associated withcentral processing module (SYS) 1205. The frequencies for the variousmodes of operation are indicated in Table 102I below. The Standard Modefrequency is derived from the CST[8] through a division by 2.

TABLE 102I Mode Frequency (khz) Shut Down 0 Low Power 0 Standard 262.144

A digital supply voltage of 3.3 V±300 mV is provided to clock generatorblock 1223 via VDD which is connected to the core digital power supply.Similarly, VSS is provided over line 1223.6 and is the core digitalground.

Turning to FIG. 35, the port table is illustrated showing the portnames, whether it is an input or an output, the description as well asthe source and destination of the signals for those ports.

Referring to FIG. 12, reset circuit 1221 generates the resets forswitching power supply controller 1200. The signals received by thiscircuit and produced by the circuit are indicated within the block. Aswith the other circuits shown in FIG. 12, the arrow adjacent to thesignal name indicates whether the signal is generated by or received byreset circuit 1221.

Section 1.1 Detailed Descriptions of the DPC and Operation, AlternativeImplementations

Digital Pulse Converter wrapper 1201 can be implemented as a combinationof a custom mixed-signal circuit (DPC) and an interface wrapper ofdigital glue logic synthesized from a logic circuit descriptionexpressed in a hardware description language (HDL). In thisimplementation, digital pulse converter wrapper 1201 converts 10-bitdigital values to pulses with edges resolved to about 2 ns. As explainedin further detail below, dual-port memory block with a single write portand a single read port is provided to store 10-bit values that expresspulse start and width control, cycle skipping and bypass circuitrycontrol (for direct output control).

FIG. 13 is a block diagram showing interface signals of digital pulseconverter wrapper 1201, according to one implementation. As shown inFIG. 13, digital pulse converter wrapper 1201 has five interfaces: (a)timing control interface 1301, (b) regulation control interface 1302,(c) power regulation interface 1303, (d) sample and hold controlinterface 1304, and (e) power supply interface 1305.

Timing control interface 1301 includes 32 KHz, 50% duty cycle referenceclock signal 1301 a (FREF), reference clock bypass control signal 1301 b(BYPASS), digital phase-locked status signal 1301 c (PLOCK), count timestate bus 1301 d (CS[9:0]), and output state bus 1301 e (STATE[15:0]),which are used to provide clocks and control states for the circuitryexternal digital pulse converter wrapper 1201. In this implementation,reference clock signal 1301 a (i.e., signal FREF) is a reference clockprovided to a digital phase-locked loop (DPLL) in digital pulseconverter wrapper 1201 for frequency synthesis, and bypass controlssignal 1301 b (i.e., the BYPASS signal) is a test control signal usedfor bypassing the DPLL. PLOCK is a status signal indicating aphase-locked condition of reference clock signal 1301 a in the DPLL.Count time state bus 1301 d (i.e., bus CS[9:0]) is a 10-bit clock statebus which provides the clocks and control states to synchronize the DPC,the interface and other core circuits of switching power supplycontroller 1200. While count time state bus 1301 d collectively displaysthe time state of the DPC, individual bits of count time state bus 1301d can be used as 50% duty cycle clocks. For example, if a 31.25 KHzreference clock is used, bit CS[9] corresponds to a 256 KHz clock with a50% duty cycle, bit CS[8] corresponds to a 512 KHz clock with a 50% dutycycle. In general CS[n] corresponds to a 50% duty cycle clock withfrequency f(n,m)=2^(−(n+m)) f₀, where nε{0, 1, . . . , 9} and mε{0, 1, .. . , 6}. Output state bus 1301 e (i.e., bus STATE[15:0]) is a 16-bitstate bus which displays the internal state of switch control buses 1303a and 1303 b (described below) prior to passing through the directcontrol logic used by regulation control interface 1303 to force theoutput signals of switch control bus 1303 a (i.e., HIGHFET) and switchcontrol bus (i.e., LOWFET) to specific states. Bus 1301 e signals whenwrite operations can occur to the DPC core circuit.

Regulation control interface 1302, which provides access to thedual-port memory in digital pulse converter wrapper 1201, includesmemory write data bus 1302 a (DWI[9:0]), memory address bus 1302 b(ADW[7:0]), memory read data bus 1302 c (DWO[9:0]), write-enable signal1302 d (WE) and read-enable signal 1302 e (RE). Regulation controlinterface 1302 controls the offsets and pulse widths for the variouspulses in power regulation interface 1303.

Power regulation interface 1303 includes switch control bus 1303 a(HIGHFET[6:0]) and switch control bus 1303 b (LOWFET[6:0]). Sample andhold control interface 1304, which controls the sampling and holding ofanalog voltages for digital conversion by the ADC used in the systemcontrol loop, includes a first sample control bus 1304 a (SMPA[6:0]), asecond sample control bus 1304 b (SMPB[6:0]), and an auxiliary controlbus 1304 c (SMPAX[3:0]). The timing for each bit in the sample buses1304 a and 1304 b is associated with each bit of the correspondingswitch bus 1303 a or switch bus 1303 b. Each of sample buses 1304 a and1034 b controls the sampling and holding of analog voltages associatedwith either the HIGHFET or LOWFET control buses for digital conversion.Sampling control bus 1304 c (i.e., auxiliary sample SMPAX[3:0]) controlsthe sampling of other analog signals needed for system monitoring andcontrol.

Power supply interface 1305 includes digital power reference 1305 c(VDD), analog power reference 1305 d (AVD), digital ground reference1305 a (VSS) and analog ground reference 1305 b (AVS). Digital power andground reference signals (i.e., VDD and VSS references) are globalsignals. In this implementation, VDD is the digital high voltage supply(3.3V±10%) connected to the core digital power supply. AVD is the analoghigh voltage supply (3.3V±10%) connected to the core analog powersupply. VSS and AVS are, respectively, the digital ground and analogground references (0V) connected to the core digital ground reference.

Section 1.1.1 First Embodiment of PWM Timing Generator

FIG. 3 shows a supply management controller of a type in accordance withthis invention. As shown in FIG. 3, a ring oscillator includes inverters301-1 through 301-15 connected in series. In an actual embodiment ofthis invention, the ring oscillator may include a larger number ofseries-connected inverters. For example, a thousand inverters can beconnected in series, with the result that the duty cycle achieved by thecontroller of this invention can be almost 100%. However, to simplifyexplanation, only 15 inverters will be illustrated in this detaileddescription. The inverters each have inherently a delay “Δ,” which isthe elapsed time between the time a signal is applied to the input leadto the inverter and the time the resulting output signal is obtained onthe output lead of each inverter. This time “Δ” is a function of thevoltage applied to the components contained within the inverter. Byvarying the voltage applied to the inverter's components, the actualdelay time Δ associated with an inverter can be varied. A typicalinverter in CMOS technology will contain a P-channel MOS device seriesconnected with an N-channel MOS device between a voltage source and areference voltage, typically system ground. Should buffers be used withan inverter an additional four transistors may be used giving eachinverter six transistors. The delay associated with the transmission ofa signal through the inverter is a function of the voltage applied tothe inverter. The higher the supply voltage applied to the inverter, theslower the transmission of the signal from the input to the output leadand the larger the delay Δ.

In the structure of FIG. 3 a crystal 302 capable of oscillating at32.768 KHz is connected by leads 303 a and 303 b to an invertingamplifier 304. Amplifier 304 provides a voltage across the crystal tocause the crystal to oscillate at 32.768 KHz, thereby causing the outputsignal from inverting amplifier 304 to oscillate at the frequency ofcrystal 302. The output signal of the ring oscillator is divided byeight in divided-by-eight circuit 305 and then sent to phase comparator306, which receives also the output signal from oscillator 302. Thus thecontrol frequency of the ring oscillator is 262.144 KHz, eight times thenormal frequency of crystal 302.

The normal operating frequency of the ring oscillator is generallyselected to be approximately the 262 Khz. The phase comparator 306senses the phase difference between the phase of the divided-by-eightoutput signal from the ring oscillator and the phase of the signal fromcrystal oscillator 302. Phase comparator 306 provides this phasedifference in an output signal on lead 306 a to Vcc control circuit 307to correct any deviation in the frequency of the ring oscillator'sdivided-by-eight output signal from the control frequency 32.768 Khzassociated with crystal oscillator 302. Should the ring oscillator be attoo low a frequency, then Vcc control circuit 307 is driven to provide ahigher voltage on lead 307 a to the inverter 301-15, thereby decreasingthe delay time associated with this inverter and thus increasing theoscillating frequency of the ring oscillator. Alternatively, if theoperating frequency of the ring oscillator is too high, then Vcc controlcircuit 307 provides a lower output voltage on output lead 307 a,thereby increasing the delay time through inverter 301-15 and thuslowering the voltage associated with this inverter.

Phase selector 308 controls the width of a pulse width modulated (PWM)output signal on output lead 310 from exclusive OR gate 309. Each of thetwo input leads to exclusive OR gate 309 is coupled by phase selector308 to a tap associated with one output lead from an inverter 301-i inthe ring oscillator. The output leads from the even-numbered inverters301 are connectable one at a time to input lead 309 a of exclusive ORgate 309. The output leads from the odd-numbered inverters 301 areconnectable one at a time to input lead 309 b of exclusive OR gate 309.The particular output leads from inverters 301-1 through 301-15 to beconnected to input leads 309 a and 309 b of inverter 309 are selecteddepending upon the requirements of the power supply being driven by thePWM signal on output lead 310 from exclusive OR gate 309.

Exclusive OR gate 309 will have a high output whenever its inputs aredifferent. This happens each time a high edge or a low edge propagatesthrough the ring oscillator past the inverters that the exclusive ORgate is attached to. Since one cycle of the ring oscillator containsboth a rising edge and a falling edge, the PWM signal observed at outputlead 310 of exclusive OR gate 309 will be twice the frequency of thering oscillator.

The normal frequency “f” of the ring oscillator (made-up of inverters301-1 through 301-15) is given by the delay time “Δ” associated witheach of the inverters. Thus if all inverters have equal delays, then thenormal frequency f=1/(2nΔ) where n is the number of inverters and Δ isthe delay time associated with each inverter. Thus the frequency isinversely proportional to the number of inverters. The period of thering oscillator is given by 1/f. Thus if the number of inverters equals1,000 and the delay associated with each inverter is 10⁻⁹ seconds, thenthe frequency is 500 Khz and the period is 2 microseconds.

The various pulse widths which the system is capable of achieving aredetermined by the taps on the output leads of the inverters which areconnected to the input leads 309 a and 309 b of exclusive OR gate 309.

Section 1.1.2 Second Embodiment of the PWM Timing Generator

FIG. 4 shows an alternative embodiment of the invention employingcounters and comparators to generate a pulse width modulated signal. Afive bit counter 41 (although a different number of bits can be used ifdesired) counts from 0 to 31 driven by a 16.7772 MHz signal. Theinstantaneous count from 5 bit counter 41 is sent on 5-bit bus 42 tocomparators 43 a and 43 b, each of which compares the count to areference count stored in it. Digital comparator 43 a will store onecount determined by the signals on phase select bus 44 a and digitalcomparator 43 b will store a second count determined by the signals onphase select bus 44 b. The signals on phase select bus 44 a and phaseselect bus 44 b are determined by external circuitry which measures thevoltage on the load capacitor and the current into the load capacitorand compares the voltage and current to reference values to determinethe extent to which the charge on the load capacitor must bereplenished. This is accomplished by switching power supply controller1200, using in particular sample and hold circuit 1207, analog todigital converter 1206 and regulation control module (REG) 1204. Thesecircuits, as well as their operation and the operation of the system aredescribed in detail below. The output signal from digital comparator 43a toggles a D flip-flop 45 a and the output signal from digitalcomparator 43 b toggles a D flip-flop 45 b. The inverting output signalfrom D flip-flop 45 a is transferred on lead 47 a to the input lead of adelay line 48 a. The delay line has a length determined by bits 4-0 soas to correspond to the time taken to drive the 5-bit counter to theparticular value which causes digital comparator 43 a to toggleflip-flop 45 a. The output signal from the delay line is transferred onlead 49 a to one input lead of exclusive OR gate 49.

5-bit counter 41 continues counting after finding a match in digitalcomparator 43 a until another match is found in digital comparator 43 b.The particular value of the count in digital comparator 43 b is set bybits 9-5 derived from phase select-red bus 44 b as described below. Thematch results in input signal being sent from digital comparator 43 b toD flip-flop 45 b. D flip-flop 45 b then produces an output signal onlead 47 b, which is transmitted to the input lead of delay line 48 b.Delay line 48 b then produces a high level output signal on lead 49 b toexclusive OR-gate 49. During the time that the output signal from delayline 48 a is high and the output signal from delay line 48 b is low, apulse width modulated signal will be produced by exclusive OR gate 49 onoutput lead 49 c. When however, the output signals on output leads 49 aand 49 b are the same level, exclusive OR gate 49 will produce a lowlevel output signal on output lead 49 c. Thus the output signal fromexclusive OR gate 49 is phase-modulated in response to the signals onphase select-green and phase select-red buses 44 a and 44 b,respectively.

Selecting 16.7772 MHz to drive 5-bit counter 41 causes it to cyclethrough its complete count 524,287½ times per second. In other words,the output frequency of the signal on lead 49 c from circuitry FIG. 4 is262,144 KHz. FIG. 5 illustrates waveforms of a string of 13-seriesconnected inverters. The top curve represents the input signal to thefirst inverter. The second curve represents the output signal from thesecond inverter. The third curve represents the output signal from thefourth inverter and the fifth, sixth, seventh and eighth curvesrepresent the output signals from the sixth, eighth, tenth and twelfthinverters, respectively. The bottom curve represents the pulse widthmodulated output signal from the circuit as shown in FIG. 3 or FIG. 4.Note that, in this example, the output signal is controlled by thesignals input to the series-connected inverters and the signal outputfrom the tenth inverter. Note that the output signal is off when thesignal input to the series-connected inverters and the signal outputfrom the tenth inverter are the same amplitude and is on when these twosignals are of complementary amplitudes. In one embodiment, discussedbelow, where the PWM signal drives a main switch of a DC/DC converter,the output current is sampled when the switch is on as shown by thedashed line and the output voltage is sampled when the switch is off asshown by the dashed line at the left hand portion of the curves. Theoutput frequency of the pulse width modulated signal is 524,288 Hz whilethe frequency of change of the signals from the series-connectedinverters is half this frequency over 262,144 Hz.

Section 1.1.3 Discussion of Counter+Comparators Approach—Without DelayLines—Optimization Techniques in the Implementation

Digital Pulse Converter wrapper 1201 may have various implementationsand include various types of interfaces to perform its functions (e.g.,pulse width modulation), as described herein. For example, pulse widthmodulation may be performed utilizing 1) a low frequency digital phaselocked loop (DPLL) voltage controlled oscillator (VCO) with inverterstages and taps (e.g., as described in reference to FIG. 3), 2) a highfrequency DPLL with a counter combined with a digital comparator, or 3)a DPLL combined with a content addressable memory (CAM) to generate therequired pulse width modulated signals.

FIG. 36 shows an exemplary pulse width modulation implementation, whichincludes a pulse width generator (PWG) 2300.4 and a sequencer 2302.4,for digital pulse converter 1201. As illustrated herein, this exemplarypulse width modulation implementation is directed to a high frequencyDPLL with a counter combined with a digital comparator (i.e., example 2as listed above), with one DPLL/counter/comparator combination for eachpower supply to be regulated.

PWG 2300.4 receives a 10-bit trigger (TRG[9:0]) signal and an 8-bit tag(TAG[7:0]) signal from sequencer 2302.4, along with a reference clock(FREF) and a reset (RST) signal. PWG 2300.4 generates a lock (PLOCK)signal, a sequencer clock (SCLK) signal, a pulse width modulation(PWM[7:0]) signal, and a clock (CLK[m:n]) signal. Sequencer 2302.4generates the trigger (TRG[9:0]) signal and the tag (TAG[7:0]) signaland receives the global reset RST signal and a data (DATA[p:0]) signalsfrom a control logic block such as regulation control module (REG) 1204.Sequencer 2302.4 also receives or provides control (CNTL[q:0]) signals.

The tag signal allows the start and stop times of different PWM signaloutputs to occur at the same trigger value (i.e., time). This allows thePWN start and stop times to slide through each other (i.e., varyindependently in time relative to each other) during normal operation.As an example of a sequence control, the trigger signal values (n0, n1,n2, . . . , n15) correspond to the tag signal values (t0, t1, t2, . . ., t15), where 1023≧n15> . . . >n2>n1>n0≧0 (where 1023 represents a timeduration from 0 to 1023 from a counter). If t1 corresponds to the startof the PWM[3] signal and t8 corresponds to the stop of the PWM[3]signal, then the pulse width of the PWM[3] signal is PWM[3]=(n8−n1)τ,where τ≅2 ns.

FIG. 36A is an exemplary implementation of pulse width generator 2300.4of FIG. 36. FIG. 36A includes a DPLL 2304.4 that generates the locksignal and an output frequency (F_(OUT)) signal that is provided to adivider counter 2308.4 that generates the clock signal. The mostsignificant bit of the clock signal is divided by 16 (by divider 2306.4)to provide a feedback signal for comparison to the reference clock.Divider counter 2308.4 also generates a 10-bit count (CNT[9:0]) signalthat is compared to the trigger signal by a comparator 2310.4, whoseoutput is clocked out to a PWM circuit 2314.4 via a flip flop 2312.4.

PWM circuit 2314.4 also receives the tag signal and generates the PWMsignal and the sequencer clock signal. An exemplary circuitimplementation for PWM circuit 2314.4 is illustrated by a circuit2316.4.

FIG. 36B is an exemplary implementation of sequencer 2302.4 of FIG. 36.FIG. 36B includes a series of registers 2330.4 (which are separatelyreferenced as 2330.4 a through 2330.4 p) and multiplexers 2332.4 (whichare separately referenced as 2332.4 a through 2332.4 n) that are clockedby the sequencer clock (SCLK) signal and generate the trigger signal andthe tag signal. Registers 2330.4 and multiplexers 2332.4 are controlledby a controller 2334.4, which receives the data signal and receives orprovides the control signals. Controller 2334.4 operations may beperformed by regulation control module (REG) 1204 or by a separatecontroller, such as a processor or a microcontroller, to provide thecontrol and data sequencing logic.

FIG. 36C shows an exemplary feedback control system for a PWM switchingvoltage regulator without a dead zone. For example, the feedback controlsystem monitors a voltage of interest (V_(M)) and compares it to atarget voltage (V_(T)) to generate an estimated PWM stop target (i.e.,stop time) for use by controller 2334.4. FIG. 36C includes ananalog-to-digital converter (ADC) 2340.4, a subtractor 2342.4, a summer2344.4, and a register 2346.4.

The voltage (V_(M)) is digitized by ADC 2340.4 and subtracted from thevoltage (V_(T) or digital DV_(T)) by subtractor 2342.4 and summed withan output of register 2346.4 by summer 2344.4. Register 2346.4 providesa PWM stop target (PWM_(ST)) signal.

If the voltage (V_(T)) is greater than the voltage (V_(M)), then the PWMstop target signal is incremented until the voltage (V_(T)) is less thanthe voltage (V_(M)), which results in the PWM stop target (PWM_(ST))signal to decrement. Once steady state is achieved, the control loopwill continue incrementing and decrementing the PWM stop target(PWM_(ST)) signal to minimize the difference between the voltage (V_(T))and the voltage (V_(M)). This negative feedback control system relies onthe fact that an increase in the PWM stop target (PWM_(ST)) signal willcause an increase in the voltage (V_(M)). The transient response of thecontrol loop is small because the change in the PWM stop target(PWM_(ST)) signal will be, for example, only one unit (e.g., ±2 ns) fromcycle to cycle.

FIG. 36D shows an exemplary feedback control system for a PWM switchingvoltage regulator with a dead zone. For example, the feedback controlsystem monitors a voltage of interest (V_(M)) and compares it to atarget voltage (V_(T)) to generate an estimated PWM stop target (i.e.,stop time) for use by controller 2334.4.

FIG. 36D includes converter (ADC) 2340.4, subtractor 2342.4, summer2344.4, register 2346.4, comparators 2348.4, and a logic gate (OR)2352.4.

The voltage (V_(M)) is digitized by ADC 2340.4 and subtracted from thevoltage (V_(T) or digital DV_(T)) by subtractor 2342.4, with the resultcompared to a positive and negative dead zone target by comparators2348.4 and 2350.4. The output of comparators 2348.4 and 2350.4 isprovided to logic gate 2352.4, whose output along with comparator 2350.4is summed by summer 2344.4 with an output of register 2346.4. Register2346.4 provides a PWM stop target (PWM_(ST)) signal.

If the difference (i.e., the voltage (V_(T)) minus the voltage (V_(M)))is less than the positive dead zone target and greater than the negativedead zone target, the PWM stop target (PWM_(ST)) signal remainsconstant. Otherwise, the PWM stop target (PWM_(ST)) signal isincremented or decremented, as required.

Section 1.1.4 Discussion of PLL/RO Using CAM, Optimization Techniques inCAM Implementation

FIG. 7 shows the output waveforms from the eleventh series-connectedinverters shown in FIG. 6. In FIG. 6 the series-connected inverters areshown having pass transistors connected from the output of each of theinverters. For the pass transistors connecting the output signals fromthe odd numbered inverters to input lead 63 a of exclusive OR-gate 63,each pass transistor is driven by a signal labeled as A, B, C, D, E, orF. Similarly, for pass transistors connecting the output signals fromthe even-numbered inverters to leads 63 b to exclusive OR gate 63, eachpass transistor is driven by a signal labeled G, H, I, J, or K. Thepulse width modulated output signal from exclusive OR gate 63 istransmitted on output lead 63 c to the load capacitor of the particularcircuit being powered by the structure shown in FIG. 6. The particularcombination of pass transistors to be turned on determines the width ofthe pulse width modulated signal output on lead 53 c from exclusive ORgate 63. Turning to FIG. 7, one can see the waveform from the inverters1 through 11. The waveform from inverter 11 is, of course, fed back inFIG. 6 to the input lead of inverter 1.

FIG. 7 and FIG. 8 show waveforms for the output signals from each of theinverters 1 through 11 in FIG. 6. FIG. 7 shows the output signals forthese inverters as taken straight from each inverter. FIG. 8 shows theoutput signals from inverters 1, 3, 5, 7, 9 and 11 taken straight fromthe output lead of each inverter while curves 2, 4, 6, 8 and 10 show thecomplement of the output signal taken from inverters 2, 4, 6, 8 and 10.FIG. 9 shows the pulse width for the various combinations of signalsapplied to input leads 63 a and 63 b of exclusive OR gate 63,respectively, throughout the pulse width of the pulse on output lead 63c from exclusive OR gate 63 is shown in the columns labeled pulse width.As can be seen from FIG. 9, only one combination of output signals fromthe various inverters is required to yield the five possible differentpulse widths obtainable using the signals directly from the inverters.Thus the pulse widths 10, 8, 6, 4 and 2 are obtained from using on inputlead 63 a of exclusive OR gate 63 the output signal from inverter 1activated by pass transistor A together with one of the output signalsfrom inverters 2, 4, 6, 8 and 10 activated by pass transistors G, H, I,J or K. Thus pulse widths of 10 delays, 8 delays, 6 delays, 4 delays and2 delays are obtained using these combinations. The only othercombination of output signals is that which uses the output signal forinverter 11 together with the output signals from inverters 2, 4, 6, 8and 10, which again yields pulse widths of 2, 4, 6, 8 and 10 delays.

If output signals from the even-numbered inverters are inverted, thenpulse widths of 1, 3, 5, 7, and 9 delays can be obtained by combining anexclusive OR gate 63 the output signal from inverter 1 with the outputsignal from inverters 2, 4, 6, 8 and 10, each inverted. Again, this setof pulse widths represent all possible pulse widths obtainable usinginverted output signals from the even-numbered inverters together withthe output signal from any one of the odd-numbered inverters.

An alternative implementation for performing pulse width modulation bydigital pulse converter 1201 includes a DPLL combined with a contentaddressable memory (CAM) to generate the required pulse width modulatedsignals (i.e., example 3 as listed above).

Digital Pulse Converter (DPC) 1201 is a low power custom mixed-signalmacro. In general, the input and output signals of DPC 1201 are digital,however separate analog power and ground signals are provided to supplyan internal digital phase lock loop (DPLL) circuit used for frequencysynthesis.

DPC 1201 synthesizes a reference clock (32.768 khz) to produce a varietyof pulses with pulse widths based on a DPC frame which starts with acount of 0 and ends with a terminal count of 1023, as illustrated in thetiming diagram shown in FIG. 37. In one implementation the DPLLgenerates a frame clock of 524.288 KHz, yielding a frame time of 1.907μSec. The minimum pulse width is represented by a count difference of 0(i.e., 0% duty cycle) and the maximum pulse width by a count differenceof 1024 (i.e., 100% duty cycle).

The rising and falling pulse edges have a minimum resolution of 1 count,which corresponds to a real time difference of ˜1.863 ns. The pulses areused to control chip I/O output drivers within NFET driver module 1202for external power regulation. The pulses are also used to control chipI/O input drivers within sample and hold module (SHM) 1207, with sampleand hold circuits for analog to digital conversion using an on chipanalog to digital converter (ADC) found within analog to digitalconverter 1206.

DPC 1201 also generates other output signals used by other chipcircuitry, for example real time clock states and synchronization pulsesfor regulation control module (REG) 1204, and source clocks for a ClockGeneration and Enable (CKGEN) macro 1223. DPC 1201 generates pulses asshown in FIG. 37, where one DPC frame corresponds to ˜1.907 us in thetiming diagram.

Each pulse (i.e., a PFET pulse 2410.4, a SFET pulse 2412.4, a SMPA pulse2414.4, and a SMPB pulse 2416.4 associated respectively with a PFETsignal 2402.4, a SFET signal 2404.4, a SMPA signal 2406.4, and a SMPBsignal 2408.4) generated by DPC 1201 is represented as a pair of 10-bitGrey coded numbers, which are presented to DPC 1201 through theinterface of Regulation control module (REG). The Grey coded numbers aredesignated as Primitive numbers or simply Primitives, with a pair ofPrimitives associated with each pulse.

The first Primitive number (i.e., a PFTS, a SFTS, a SPAS, and a SPBS inFIG. 37) in each pair indicates the offset of the leading edge of thepulse from a count of 0 and can be any integer from 0 to 1023. Thesecond Primitive number (i.e., a PFTR, a SFTR, a SPAR, and a SPBR inFIG. 37) in each pair indicates the offset of the trailing edge of thepulse from a count of 0 and can also be any integer from 0 to 1023. Theletter “S” on the end of signal parameters for the first Primitivenumbers (PFTS, SFTS, SPAS, and SPBS) stands for SET and the letter “R”on the end of signal parameters for the second Primitive numbers (PFTR,SFTR, SPAR, and SPBR) stands for RESET.

PFET pulse 2410.4 and SFET pulse 2412.4, shown along the ordinate of thetiming diagram in FIG. 37, control Primary and Secondary NFET drivers inNFET driver module 1202 and SMPA pulse 2406.4 and SMPB pulse 2416.4control the input sample and hold circuits in SHM 1207. The pulses shownin the diagram can be represented by 8 independent 10-bit Primitives.

The eight independent Primitives are designated as follows: PFTS, PFTR,SFTS, SFTR, SPAS, SPAR, SPBS and SPBR.

These primitive numbers (i.e., PFTS, PFTR, SFTS, SFTR, SPAS, SPAR, SPBSand SPBR) are stored in a special purpose, dual port content addressablememory (CAM) device in DPC 1201 and the notation for these numbers aredescribed in detail for port descriptions below. The timing diagram inFIG. 37 shows the Primitive number set for a single pulse channelcontrolled by DPC 1201, however DPC 1201 can provide a number ofindependent pulse channels (e.g., seven independent pulse channels forexternal PWM-switching power conversion and an eighth independentauxiliary pulse channel which can be used for synchronization ofinternal or external circuitry).

If DPC 1201 provides eight independent pulse channels, then a busnotation for PFET signal 2402.4, SFET signal 2404.4, SMPA signal 2406.4,and SMPB signal 2408.4 may be used to designate these independentchannels as a PFET[7:0] signals 2454.4, a SFET[7:0] signals 2452.4, aSMPA[7:0] signals 2450.4, and a SMPB[7:0] signals 2448.4, respectively,as discussed below in reference to FIG. 37A. PFET[7], SFET[7], SMPA[7]and SMPB[7] refer to the auxiliary pulse channel, and the PFET[6:0]signals 2454.4 and SFET[6:0] signals 2452.4 refer to pulses whichcontrol drivers in NFET driver module 1202 to turn on and off externalpower FETs used for power regulation. The SMPA[6:0] signals 2450.4 andSMPB[6:0] signals 2448.4 refer to pulses which control the input sampleand hold circuits in SHM 1207 for digitizing external analog voltages.It should be noted that the timing diagram shown in FIG. 37 illustratesa break before make switching algorithm between PFET signal 2402 andSFET signal 2404.4 (i.e., between the Primitive number PFTR and thePrimitive number SFTS), which is generally needed for efficient powerregulation.

DPC 1201 for this implementation has five interfaces to communicate withfive corresponding portions of switching power supply controller 1200(i.e., IVS 1209, CKGEN 1223, regulation control module (REG) 1204, NFETdriver module 1202, and SHM 1207). FIG. 37A illustrates this exemplaryinterface implementation for DPC 1201. The interface with IVS 1209includes the FREF signal 2420.4.

Additionally, the interface with IVS 1209 includes power and groundsignals, including a VDD signal 2464.4, an AVD signal 2466.4, a VSSsignal 2442.4, and an AVS signal 2444.4. The digital power and ground,VDD signal 2464.4 (e.g., 3.3 V) and VSS signal 2442.4, respectively, canbe treated as global signals, whereas the analog power and ground, AVDsignal 2466.4 (e.g., 3.3 V) and AVS signal 2444.4, respectively, aregenerally not treated as global signals.

FREF signal 2420.4 is a reference clock provided to the DPLL in DPC 1201and has a frequency and a duty cycle of about 32.768 kHz and 50%,respectively. PLOCK signal 2462.4 is a signal which is asserted (i.e.,transitions to a logical high or HIGH state) and remains asserted (i.e.,remains at a logical high) after the DPLL achieves and remains in aphase lock condition. Otherwise PLOCK is not asserted (i.e., a logicallow or LOW state). AUX signal 2446.4 is an auxiliary signal port that isused for synchronizing circuitry external to switching power supplycontroller 1200.

The interface with Clock Generator and Enable (CKGEN) 1223 includesvarious signals, including a CST[9:0] signal 2458.4, a PLLCK signal2460.4, and a PLOCK signal 2462.4. CST[9:0] signal 2458.4 is a 10-bitGrey coded clock state bus which provides the clocks and control statesto synchronize DPC 1201, regulation control module (REG) 1204, and theCKGEN macro 1223. In a standard operating mode, with SSC signal 2424.4set LOW and FREF signal 2420.4 set to 32.768 kHz, exemplary frequenciesfor the CST[9:0] signals 2458.4 are given in Table 1. PLLCK signal2460.4 is the DPLL output clock which has a frequency and a duty cycleof 524.288 kHz and 50% respectively, assuming a reference frequency forFREF signal 2420.4 of 32.768 kHz. The same frequency for the last twomost significant bits of CST[9:0] signals 2458.4 is an artifact of a10-bit Grey counter generating these frequencies. The MSB (mostsignificant bit) and NMSB (next most significant bit), (CST[9] andCST[8], respectively, of CST[9:0] signal 2458.4) have a quadrature phaserelationship to each other.

TABLE 1 CST Bit Frequency kHz CST[0] 134,217.728 CST[1] 67,108.864CST[2] 33,554.432 CST[3] 16,777.216 CST[4] 8,388.608 CST[5] 4,194.304CST[6] 2,097.152 CST[7] 1,048.576 CST[8] 524.288 CST[9] 524.288

The interface with regulation control module (REG) 1204 includes varioussignals, including PD_OUT[1:0] signals 2426.4, SET[28:0] signals 2438.4,RST[28:0] signals 2440.4, ENBL[21:0] signals 2436.4, DWI[19:0] signals2428.4, DRO[19:0] signals 2456.4, and ADW[4:0] signals 2430.4, which areinput buses, and the CST[9:0] signals 2458.4, which is an output bus.Additionally, the interface includes an input signal WE 2432.4, an inputsignal RE 2434.4, and an output signal PLOCK 2462.4.

The interface with the regulation control module (REG) 1204 is used togenerate the various pulses, such as PFET pulse 2410.4 and SFET pulse2412.4. The DWI[19:0] signals 2428.4, ADW[4:0] signals 2430.4, DRO[19:0]signals 2456.4, and input signal WE 2432.4 and input signal RE 2434.4transfer data and control the read/write ports of the dual port CAM,discussed in further detail herein. The read port for the CAM is withinDPC 1201.

More specifically, the PD_OUT[1:0] signals 2426.4 are a 2 bit bus whichcontrols the mode of operation for DPC 1201. The modes of operation aredesignated as Standard mode, Low Power mode, and Shut Down mode, assummarized in Table 2.

When PD_OUT[1:0] signals 2426.4 are asserting the Shut Down mode, theDPLL is powered down, the CAM is in standby mode, and the rest of thedigital blocks in DPC 1201 are in low power states. When PD_OUT[11:0]signals 2426.4 are asserting the Low Power mode, the DPLL is powered upand phase locked at its normal operating frequency (536,870.912 kHz),the DPLL output is divided by 16 to produce the LSB of CST[9:0] signal2458.4 (i.e., the frequency of the CST LSB is 33,554.432 kHz), the CAMis in standby mode, and the remaining blocks of DPC 1201 are in lowpower states. When PD_OUT[1:0] signals 2426.4 are asserting the Standardmode, the DPLL is operating normally, the DPLL output is divided by 4 toproduce the LSB of CST[9:0] signals 2458.4 (i.e., the frequency of theCST LSB is 134,217.728 kHz), the CAM is powered up and operatingnormally and the rest of the DPC blocks are also powered up and areoperating normally.

TABLE 2 Exemplary Modes PD OUT[1] PD OUT[0] MODE 0 0 Shut Down 0 1TBD/Spare 1 0 Low Power 1 1 Standard

The SET[31:0] signals 2438.4 are a 32-bit control bus which is used toindependently set each output bit of DPC 1201. For example, when SET[0]of the SET[31:0] signals 2438.4 is HIGH, SMPA[0] of the SMPA[7:0]signals 2450.4 is set HIGH, and when SET[1] of the SET[31:0] signals2438.4 is HIGH, PFET[0] of the PFET[7:0] signals 2454.4 is set HIGH.Table 3 illustrates exemplary associations between SET[31:0] signal2438.4, the Primitive numbers, and output signals from DPC 1201.

TABLE 3 Set Bits Primitive Numbers DPC Macro Outputs SET[0]˜SET[3]SPAS[0], PFTS[0], SPBS[0], SFTS[0] SMPA[0], PFET[0], SMPB[0], SFET [0]SET[4]˜SET[7] SPAS[1], PFTS[1], SPBS[1], SFTS[1] SMPA[1], PFET[1],SMPB[1], SFET [1] SET[8]˜SET[11] SPAS[2], PFTS[2], SPBS[2], SFTS[2]SMPA[2], PFET[2], SMPB[2], SFET [2] SET[12]˜SET[15] SPAS[3], PFTS[3],SPBS[3], SFTS[3] SMPA[3], PFET[3], SMPB[3], SFET [3] SET[16]˜SET[19]SPAS[4], PFTS[4], SPBS[4], SFTS[4] SMPA[4], PFET[4], SMPB[4], SFET [4]SET[20]˜SET[23] SPAS[5], PFTS[5], SPBS[5], SFTS[5] SMPA[5], PFET[5],SMPB[5], SFET [5] SET[24]˜SET[27] SPAS[6], PFTS[6], SPBS[6], SFTS[6]SMPA[6], PFET[6], SMPB[6], SFET [6] SET[28]˜SET[31] SPAS[7], PFTS[7],SPBS[7], SFTS[7] SMPA[7], PFET[7], SMPB[7], SFET [7]

The RST[31:0] signals 2440.4 are a 32-bit control bus which is used toindependently reset each output bit of DPC 1201. For example, whenRST[0] of the RST[31:0] signals 2440.4 is HIGH, the SMPA[0] output ofthe SMPA[7:0] signals 2450.4 is reset LOW, and when RST[1] of theRST[31:0] signals 2440.4 is HIGH, the PFET[0] output of the PFET [6:0]signals 2454.4 is reset LOW. Table 4 illustrates exemplary associationsbetween RST[31:0] signal 2440.4, the Primitive numbers, and outputsignals from DPC 1201.

TABLE 4 Reset Bits Primitive Numbers DPC Macro Outputs RST[0]˜RST[3]SPAR[0], PFTR[0], SPBR[0], SFTR SMPA[0], PFET[0], SMPB[0], SFET[0] [0]RST[4]˜RST[7] SPAR[1], PFTR[1], SPBR[1], SFTR SMPA[1], PFET[1], SMPB[1],SFET[1] [1] RST[8]˜RST[11] SPAR[2], PFTR[2], SPBR[2], SFTR SMPA[2],PFET[2], SMPB[2], SFET[2] [2] RST[12]˜RST[15] SPAR[3], PFTR[3], SPBR[3],SFTR SMPA[3], PFET[3], SMPB[3], SFET[3] [3] RST[16]˜RST[19] SPAR[4],PFTR[4], SPBR[4], SFTR SMPA[4], PFET[4], SMPB[4], SFET[4] [4]RST[20]˜RST[23] SPAR[5], PFTR[5], SPBR[5], SFTR SMPA[5], PFET[5],SMPB[5], SFET[5] [5] RST[24]˜RST[27] SPAR[6], PFTR[6], SPBR[6], SFTRSMPA[6], PFET[6], SMPB[6], SFET[6] [6] RST[28]˜RST[31] SPAR[7], PFTR[7],SPBR[7], SFTR SMPA[7], PFET[7], SMPB[7], SFET[7] [7]

The ENBL[23:0] signals 2436.4 are a 24-bit CAM enable bus used toindependently enable the CAM match ports. The ENBL[3n] bits ofENBL[23:0] signal 2436.4 enable the SMPA[n] bits of the SMPA[7:0]signals 2450.4 and the SMPB[n] bits of the SMPB[7:0] signals 2448.4 forCAM match outputs, where nε{0, 1, 2, . . . , 7}. The ENBL[3n+1] bits ofthe ENBL[23:0] signals 2436.4 enable the PFET[n] bits of the PFET[7:0]signals 2454.4 for the CAM match outputs and the ENBL[3n+2] bits of theENBL[23:0] signals 2436.4 enable the SFET[n] bits of the SFET[7:0]signals 2452.4 for the CAM match outputs, where nε{0, 1, 2, . . . , 7}.The CAM ENBL bus (i.e., the ENBL[23:0] signals 2436.4) and outputcorrespondence for an exemplary implementation is given in Table 5.

TABLE 5 ENBL Bus Bits DPC Macro Outputs ENBL[0] SMPA[0], SMPB[0] ENBL[1]PFET[0] ENBL[2] SFET[0] ENBL[3] SMPA[1], SMPB[1] ENBL[4] PFET[1] ENBL[5]SFET[1] ENBL[6] SMPA[2], SMPB[2] ENBL[7] PFET[2] ENBL[8] SFET[2] ENBL[9]SMPA[3], SMPB[3] ENBL[10] PFET[3] ENBL[11] SFET[3] ENBL[12] SMPA[4],SMPB[4] ENBL[13] PFET[4] ENBL[14] SFET[4] ENBL[15] SMPA[5], SMPB[5]ENBL[16] PFET[5] ENBL[17] SFET[5] ENBL[18] SMPA[6], SMPB[6] ENBL[19]PFET[6] ENBL[20] SFET[6] ENBL[21] SMPA[7], SMPB[7] ENBL[22] PFET[7]ENBL[23] SFET[7]

The ENBL bits of the ENBL[23:0] signals 2436.4 are active HIGH. Toenable specific CAM match ports, the corresponding enable bit is setHIGH. The ENBL[23:0] signals 2436.4 affects only the read port of theCAM and the Primitive numbers can be read from and written to the CAMthrough the read/write port, as described in further detail below. Thisfeature allows the CAM to be safely updated without causing aninadvertent match during a CAM update. In addition, the ENBL[23:0]signals 2436.4 allow the capability for pulse skipping during the normaloperation of the CAM.

The DWI[19:0] signals 2428.4 are a 20-bit write data bus for theread/write CAM port and it is used for writing Grey coded words to theCAM. The DWI[19:0] bus 2428.4 writes to the CAM are controlled by theADW address bus (i.e., the ADW[4:0] signals 2430.4) and the WE signal2432.4. The DWI[9:0] bits of DWI[19:0] signal 2428.4 are allocated forthe bank zero of the CAM and the DWI[19:10] bits of DWI[19:0] signal2428.4 are allocated for bank one of the CAM.

The ADW[4:0] signals 2430.4 are a 5-bit address bus used to address asingle (20-bit) word in the CAM for reading or writing. For ease ofimplementation and in accordance with one embodiment, the CAM is splitinto two banks, as described in Table 6. The ADW[4:0] signals 2430.4simultaneously addresses one 10-bit word out of 22 words in each CAMbank. For example, ADW[0] of the ADW[4:0] signals 2430.4 corresponds tothe Primitive SPBS[0] in the CAM bank zero and the Primitive SPBR[0] inthe CAM bank one.

TABLE 6 Read/Write CAM Bank Zero CAM Bank One Port ADW Primitive NumbersPrimitive Numbers ADW[0] SPBS[0] SPBR[0] ADW[1] PFTS[0] PFTR[0] ADW[2]SFTS[0] SFTR[0] ADW[3] SPAS[0] SPAR[0] ADW[4] SPBS[1] SPBR[1] ADW[5]PFTS[1] PFTR[1] ADW[6] SFTS[1] SFTR[1] ADW[7] SPAS[1] SPAR[1] ADW[8]SPBS[2] SPBR[2] ADW[9] PFTS[2] PFTR[2] ADW[10] SFTS[2] SFTR[2] ADW[11]SPAS[2] SPAR[2] ADW[12] SPBS[3] SPBR[3] ADW[13] PFTS[3] PFTR[3] ADW[14]SFTS[3] SFTR[3] ADW[15] SPAS[3] SPAR[3] ADW[16] SPBS[4] SPBR[4] ADW[17]PFTS[4] PFTR[4] ADW[18] SFTS[4] SFTR[4] ADW[19] SPAS[4] SPAR[4] ADW[20]SPBS[5] SPBR[5] ADW[21] PFTS[5] PFTR[5] ADW[22] SFTS[5] SFTR[5] ADW[23]SPAS[5] SPAR[5] ADW[24] SPBS[6] SPBR[6] ADW[25] PFTS[6] PFTR[6] ADW[26]SFTS[6] SFTR[6] ADW[27] SPAS[6] SPAR[6] ADW[28] SPBS[7] SPBR[7] ADW[29]PFTS[7] PFTR[7] ADW[30] SFTS[7] SFTR[7] ADW[31] SPAS[7] SPAR[7]

The DRO[19:0] signals 2456.4 are a 20-bit read data bus for theread/write CAM port and it is used for reading Grey coded words from theCAM. The DRO[19:0] bus 2456.4 reads from the CAM are controlled by theADW address bus (i.e., the ADW[4:0] signals 2430.4) and the RE signal2434.4. The DRO[9:0] bits of DRO[19:0] signal 2456.4 are allocated forthe bank zero of the CAM and the DRO[19:10] bits of DRO[19:0] signal2456.4 are allocated for bank one of the CAM.

The PFET[6:0] signals 2454.4 are a 7-bit bus which provides the primarypower FET pulses (i.e., pulses, such as PFET pulse 2410.4) to NFETdriver module 1202. The SFET[6:0] signals 2452.4 are a 7-bit bus whichprovides the secondary power FET pulses (i.e., pulses, such as SFETpulse 2412.4) to NFET driver module 1202.

The WE signal 2432.4 is the write enable control signal for theread/write CAM port. When WE signal 2432.4 toggles HIGH, a 10-bit wordis written to each bank of the CAM at the address specified by theADW[4:0] signals 2430.4. The RE signal 2434.4 is the read enable controlsignal for the read/write CAM port. When the RE signal 2434.4 togglesHIGH, a 10-bit word is read from each bank of the CAM at the addressspecified by the ADW[4:0] signals 2430.4.

The interface for NFET driver module 1202 includes various signals,including the PFET[6:0] signals 2454.4 and SFET[6:0] signals 2452.4. Asdiscussed above, a single PFET (primary FET) and SFET (secondary FET)pulse channel is shown in the above timing diagram (FIG. 37).

The interface for SHM 1207 includes various signals, including theSMPA[6:0] signals 2450.4 and the SMPB[6:0] signals 2448.4. Thisinterface is used to control the sampling and holding of analog voltagesfor digital conversion by analog to digital converter 1206. As describedabove in accordance with one embodiment, the sample pulses SMPA (e.g.,SMPA pulse 2414.4) and SMPB (e.g., SMPB pulse 2416.4) are independentfrom the PFET (Primary FET) and SFET (Secondary FET) pulses. Either ofthe SMPA[6:0] signals 2450.4 or the SMPB[6:0] signals 2448.4 can be usedto control the sampling (and holding) of analog voltages in SHM 1207 foranalog to digital converter 1206. The remaining signals shown in FIG.37A and described in Table 8 include a bypass signal 2422.4 and the SSCsignal 2424.4. Bypass signal 2422.4 is a test control signal used forbypassing the DPLL. When bypass signal 2422.4 is held HIGH, FREF signal2420.4 bypasses the DPLL, but when bypass signal 2422.4 is held LOW,FREF signal 2420.4 is used for frequency synthesis. SSC signal 2424.4 isa control signal that activates spread spectrum clocking. Spreadspectrum clocking is activated when SSC signal 2424.4 is HIGH; otherwisespread spectrum clocking is disabled. The spread spectrum clockingscheme implemented in DPC 1201 can be either up or down frequencyspreading where the DPC frame frequency (e.g., 524.288 kHz withoutspread spectrum clocking) deviates from its fundamental frequency byapproximately 0.5% with a modulation period of approximately 22 us.

TABLE 8 Descriptive Summary of Exemplary Interface Signals Desti- PortName Type Description Source nation FREF Input 32.768 khz IVS 1209Reference Clock BYPASS Input Reference Clock REG 1204 Bypass Control SSCInput Spread Spectrum REG 1204 Clock Control PLOCK Output DPLL Lock,CKGEN Active HIGH 1223 PD_OUT[1:0] Input Power REG 1204 ManagementControl Bus PLLCK Output DPLL Output CKGEN Clock 1223 CST[9:0] OutputCount Time State CKGEN Bus 1223 REG 1204 ENBL [21:0] Input CAM SectionREG 1204 Enable Bus DWI[19:0] Input CAM Read/Write REG 1204 Port DataBus ADW [4:0] Input CAM Read/Write REG 1204 Port Address Bus WE InputCAM Read/Write REG 1204 Port Write Enable RE Input CAM Read/Write REG1204 Port Read Enable DRO[19:0] Output CAM Read/Write REG 1204 Port DataBus SET[28:0] Input Pulse Set Bus REG 1204 RST[28:0] Input Pulse ResetBus REG 1204 PFET[6:0] Output Primary NFET 1202 FET Control BusSFET[6:0] Output Secondary NFET 1202 FET Control Bus SMPA[6:0] OutputSample A Control SHM 1207 Bus SMPB[6:0] Output Sample B Control SHM 1207Bus AUX Output Auxiliary Pulse IVS 1209 Control Port VDD Power DigitalPower IVS 1209 AVD Power Analog Power IVS 1209 VSS Power Digital GroundIVS 1209 AVS Power Analog Ground IVS 1209

FIG. 37B illustrates one circuit implementation for DPC 1201 andincludes a DPLL 2480.4, a spreader divider 2482.4, a Grey counter2484.4, and a CAM module 2486.4. CAM module 2486.4 includes a modifiedCAM for generating PFET[6:0] signal 2454.4, SFET[6:0] signal 2452.4,SMPA[6:0] signal 2450.4, SMPB[6:0] signal 2448.4, and AUX signal 2446.4.FIG. 37C illustrates one exemplary implementation for CAM module 2486.4.

CAM module 2486.4 stores, for example, 64 words (i.e., Primitives, whichwere discussed above), with 10-bits per word and with an address readport 2502.4 (labeled ADR[63:0]) of CAM module 2486.4 not encoded.Address read port 2502.4 provides 64 address signals, referencedcollectively as ADR[63:0] signals 2508.4. The even ADR bits (ADR[0],ADR[2], . . . , ADR[62]) of ADR[63:0] signals 2508.4 are associated withCAM bank zero and the odd ADR bits (ADR[1], ADR[3], . . . , ADR[63]) ofADR[63:0] signals 2508.4 are associated with the CAM bank one, aspreviously discussed (e.g., in reference to ADW[4:0] signal 2430.4).ADR[63:0] signals 2508.4 are connected to 32 RS latches, which arerepresented by an RS latch 2504.4 (FIG. 37C), through control logic2506.4, shown in a representative fashion in FIG. 37C. A more detailedexemplary implementation for control logic 2506.4 and RS latch 2504.4,associated with a single pulse channel at the output of CAM module2486.4, is shown in FIG. 37D and discussed below.

DPLL 2480.4 (FIG. 37B) is shown coupled to Grey counter 2484.4 (e.g., afree running 10-bit Grey Counter), with spreader divider 2482.4 situatedbetween DPLL 2480.4 and Grey counter 2.484.4. When SSC signal 2424.4 isenabled, spreader divider 2482.4 produces spread spectrum clocking, asdescribed above, by employing a uniform pulse swallowing technique whichvaries in frequency. Spread spectrum mode of operation is describedbelow.

Additionally, spreader divider 2482.4 provides the variable divide ratiofor toggling between the standard and low power modes, described abovefor PD_OUT[1:0] signal 2426.4. The extra division provided by spreaderdivider 2482.4 reduces the current drawn by Grey counter 2484.4 in thelow power mode. For example, as compared to a binary counter, Greycounter 2484.4 produces glitch free read operations for CAM module2486.4.

An example of operation for the implementation shown in FIG. 37B for DPC1201 starts with DPLL 2480.4, which increments Grey counter 2484.4(i.e., through spreader divider 2482.4) to produce the read data,identified by a DRI signal 2488.4, for CAM Module 2486.4. If the readdata of DRI signal 2488.4 generates a CAM match in CAM Module 2486.4,one or more of the CAM output read address lines (i.e., ADR[63:0]signals 2508.4) of CAM Module 2486.4 becomes active, which sets orresets one or more of the 32 RS latches (i.e., RS latch 2504.4) toproduce the output pulses on the PFET[7:0] signals 2454.4, SFET[7:0]signals 2452.4, SMPA[7:0] signals 2450.4 and SMPB[7:0] signals The 32 RSlatches (represented by RS latch 2504.4) are organized into eight pulsechannels. The 4 pulses within a pulse channel are completely independentas described earlier And the eight independent Primitive numbers (oneset of eight numbers for each pulse channel) are Grey coded and writtento specific address locations in the CAM of CAM module 2486.4 regulationcontrol module (REG) 1204. The address locations for the Primitivenumbers are given above in reference to ADW[4:0] signal 2430.4.

CAM Module's 2486.4 logic that is associated with one pulse channel(e.g., given by a PFET[n] signal 2512.4, an SFET[n] signal 2516.4, anSMPA[n] signal 2510.4, and an SMPB[n] signal 2514.4, where nε{0, 1, 2, .. . , 7}) is shown in detail in FIG. 37D. Eight independent CAM ADRlines (e.g., ADR[8n], ADR[8n+1], ADR[8n+2], ADR[8n+3], ADR[8n+4],ADR[8n+5], ADR[8n+6] and ADR[8n+7]) from the ADR[63:0] signals 2508.4are shown controlling four RS latches (separately referenced as RSlatches 2504.4(1) through 2504.4(4)) of RS latch 2504.4.

The control circuitry for the ENBL[23:0] signals 2436.4, which areseparately referenced as enable control logic 2506.4(1) of control logic2506.4, is also shown in detail in FIG. 37D. The ENBL[3n+1] signals andENBL[3n+2] signals, of the ENBL[23:0] signals 2436.4, control RS latches2504.4(2) and 2504.4(4) for the PFET[n] signals 2512.4 and SFET[n]signals 2516.4, respectively, and the ENBL[3n] signals of the ENBL[23:0]signals 2436.4 controls RS latches 2504.4(1) and 2504.4(3) for theSMPA[n] signals 2510.4 and SMPB[n] signals 2514.4, respectively. Table5, discussed above, provides additional details regarding the ENBL[23:0]signals 2436.4.

The control circuitry for the SET[31:0] signals 2438.4 and the RST[31:0]signals 2440.4, which are-separately referenced as set/reset controllogic 2506.4(2) of control logic 2506.4, is also shown in detail in FIG.37D. The SET[31:0] signals 2438.4 and RST[31:0] signals 2440.4 allowdirect control of RS latch 2504.4 (e.g., RS latches 2504.4(1) through2504.4(4)) at the output of CAM Module 2486.4). In the low power mode,this interface is used in conjunction with the ENBL[23:0] signals 2436.4by logic in regulation control module (REG) 1204 to generate the ADCsamples and power regulation pulses needed for operation in this mode.Using these control signals, regulation control module (REG) 1204 cantake direct control of RS latch 2504.4 to avoid hazard conditions (e.g.,preventing the set (S) and reset (R) inputs to each latch from beingactive simultaneously).

As shown in FIG. 37C, a CAM 2494.4 is a dual-port memory device with oneread/write port 2496.4 (associated with the DWI[19:0] signals 2428.4,ADW[4:0] signals 2430.4, WE signal 2432.4, RE signal 2434.4, andDRO[19:0] signals 2456.4) and one read port 2498.4 (associated with theDRI signals 2488.4 and ADR[63:0] signals 2508.4). The same Primitivenumber can be written to two or more address locations in CAM 2494.4,which will cause multiple matches on the ADR[63:0] signals 2508.4 (i.e.,the CAM read address lines) when the matching data is presented to readport 2498.4 (through the DRI signals 2488.4) of CAM 2494.4 by GreyCounter 2484.4. The multiple matches allow two or more output edges tocoincide.

Regulation control module (REG) 1204 synchronizes the writes to CAM2494.4 to avoid the generation of inadvertent glitches on the variouspulse channels. When key independent pulse edges (i.e., master edges)change from one DPC frame of DPC 1201 to the next, regulation controlmodule (REG) 1204 re-computes the dependent pulse edges (i.e., slaveedges) and updates CAM 2494.4 accordingly. The master edges correspondto the Primitives PFTS[n], PFTR[n], SFTS[n], and SFTR[n], where nε{0, 1,. . . , 7}. All other edges are either slave edges or auxiliary edges.

If a master edge is moved from its position (Grey coded count) in thecurrent DPC frame of DPC 1201 to a different position in the next frame,the slave edges related to the specific master edge are re-computed andwritten to CAM 2494.4 by regulation control module (REG) 1204 for use inthe next frame. Regulation control module (REG) 1204 provides thecomputations, Grey coding, and CAM write coordination required toperform these tasks efficiently and without hazards.

The information necessary to compute the slave edges from the masteredge information is contained within regulation control module (REG)1204, and this information does not generally change from one DPC frameof DPC 1201 to the next. For example, the slave edge informationnecessary to calculate the Primitive data SPAS and SPBS edges for eachchannel can be provided as a single 10-bit binary constant (or pulsewidth), which is used by regulation control module (REG) 1204 to computeand Grey encode a new Primitive SPAS value if a Primitive PFTS valuechanges or (more likely) to compute and Grey encode a new Primitive SPBSvalue if a given Primitive PFTR value changes.

The master edge information provided by regulation control module (REG)1204 can change from one DPC frame of DPC 1201 to the next. Regulationcontrol module (REG) 1204 provides this information as pairs of 10-bitGrey coded numbers comprised of two offsets from a DPC 1201 frameinitial count of zero. One offset corresponds to the SET input of one ofthe RS latches (i.e., RS latch 2504.4) at the output of CAM module2486.4 and the other offset corresponds to the RESET input of the RSlatch.

As mentioned above, read port 2498.4 of CAM 2494.4 (associated with DRIsignal 2488.4) is buried within DPC 1201, as illustrated in FIG. 37B andFIG. 37C. The ENBL[23:0] signals 2436.4 affects only the read port ofCAM 2494.4 and is used for the multiple functions described above. If aspecific ENBL bus bit of the ENBL[23:0] signals 2436.4 is held LOW, forexample ENBL[0] bit, the read address section corresponding to theENBL[0] bit is disabled and any data match occurring in thecorresponding disabled section (e.g., the Primitives SPAS[0], SPAR[1],SPBS[0] and SPBR[0]) will not produce a match. However, if the same dataoccurs in another read address section of CAM 2494.4, which is notdisabled, the match will occur.

Because the ENBL[23:0] signals 2436.4 only affects the read port of CAM2494.4, read/write port 2496.4 is not affected. Therefore, reads andwrites to CAM 2494.4 through read/write port 2496.4 can take placeunimpeded. This capability can be used in conjunction with the PFET[7:0]signals 2454.4 or SFET[7:0] signals 2452.4 to safely update thePrimitive numbers in CAM 2494.4 and avoid the inadvertent generation ofpulse glitches. One possible update sequence is shown in the timingdiagram of FIG. 37E for updating the Primitive SFTR[0] in CAM 2494.4.

The Primitive SFTR[0] update to CAM 2494.4 starts with the rising edgeof an SFET[0] bit of the SFET[7:0] signals 2452.4. At the rising edge ofthe SFET[0] bit, the SFTS[0] Primitive match has already occurred.Regulation control module (REG) 1204 detects the rising edge of theSFET[0] bit and disables the required CAM section of CAM 2494.4 oneregulation control module (REG) 1204 clock cycle later by setting anENBL[2] signal LOW. Note that this does not present a problem for aPFET[0] pulse of the PFET[7:0] signals 2454.4, because the rising andfalling edges of that pulse have already occurred.

After the CAM section of CAM 2494.4 is disabled, regulation controlmodule (REG) 1204 enables a CAM write by toggling WE signal 2432.4 HIGH.Because the ADW[4:0] signals 2430.4 (i.e., the read/write address portADW) has been set to address location 2, which corresponds to thePrimitive SFTR[0] as shown in Table 6, the new Primitive for SFTR[0] iswritten to the CAM address location 2 of CAM 2494.4. Regulation controlmodule (REG) 1204 then re-activates the section by setting the ENBL[2]bit HIGH and the new falling edge of the SFET[0] pulse of the SFET[7:0]signals 2452.4 occurs at the new SFTR[0] Primitive value when the CAMmatch occurs as indicated by the falling edge of the SFET[0] pulse.

Changing some Primitives has more of an effect on other Primitives thanthe example given in reference to FIG. 37E. Because of the dependenciesdiscussed herein, if the Primitive PFTR is changed, the Primitives SPBS,SFTS, and SFTR may need to be recomputed, Grey coded, and updated to CAM2494.4 by regulation control module (REG) 1204. As another example, ifthe Primitive PFTS is changed all of the Primitives SPAS, SPBS, PFTR,SFTS, and SFTR may need to be recomputed, Grey coded, and updated to CAM2494.4 by regulation control module (REG) 1204.

Because the worst case write cycle time for CAM 2494.4 is, for example,7.5 ns, the updates described in the previous sentence can be completedin 22.5 ns (i.e., 3 times 7.5 ns). To update the entire 64 words in CAM2494.4 would take approximately 240 ns. In FIG. 37, as an example, abreak before make switching algorithm for PFET signal 2402.4 and SFETsignal 2404.4 was presented. If the break time between PFET signal2402.4 and SFET signal 2404.4 were always greater than 30 ns, all of thePrimitives (eight numbers) necessary to update a pulse channel can bewritten to CAM 2494.4 during this period.

Section 1.1.5 Discussion of Combinations and Permutations of PLL/RO, DLLand Counters for Optimization of Standby Power and Die Size.

FIG. 38 shows a circuit 2600.4, which is an exemplary implementation forGrey counter 2484.4 (FIG. 37B). Circuit 2600.4 represents a 10-bit Greycounter, but could be implemented as any number of bits based upon thetechniques discussed herein. Furthermore, circuit 2600.4 can beimplemented as a low power circuit utilizing circuit techniquesdiscussed below (e.g., in reference to FIG. 38B through FIG. 38F.

Circuit 2600.4 includes a number of flip flops 2602.4, which areseparately referenced as flip flop 2602.4(1) through 2602.4(10), alongwith a number of AND gates 2604.4 and a number of exclusive OR (XOR)gates 2606.4. Circuit 2600.4 generates a 10-bit Grey count (i.e., bitsC0 through C9 in FIG. 38) that is provided to CAM module 2486.4 (FIG.37B).

FIG. 38A shows a circuit 2610.4, which is another exemplaryimplementation for Grey counter 2484.4. Circuit 2610.4 is similar toCircuit 2600.4, but utilizes NAND gates 2612.4 and NOR gates 2614.4rather than AND gates 2604.4. As shown in FIG. 38 and FIG. 38A, flipflop 2602.4(1) receives a clock (CLK) signal 2603.4 while flip flops2602.4(2) through 2602.4(10) receive a clock signal (CLK bar) 2605.4,which is the complement of clock signal 2603.4. Circuits 2600.4 and2610.4 may be implemented using low power circuit techniques discussedin the following figures.

FIG. 38B and FIG. 38C illustrate circuit implementations that mayrequire less power than conventional flip flop circuits by utilizingfewer circuit elements. FIG. 38B shows an exemplary implementation forflip flop 2602.4(1) of FIG. 38 or FIG. 38A. FIG. 38B includestransistors 2620.4 and 2624.4 (i.e., a p-type and an n-type transistor,respectively) along with inverters 2622.4, 2626.4, and 2628.4.Transistor 2620.4 receives a D input signal 2630.4 and transistors2620.4 and 2624.4 receive a clock (CLK) signal 2632.4, with inverters2626.4 and 2628.4 providing a Q signal 2634.4 and a QN (Q NOT or Q bar)signal 2636.4, respectively.

In a similar fashion, FIG. 38C shows an exemplary implementation forflip flops 2602.4(2) through 2602.4(10) of FIG. 38 or FIG. 38A. FIG. 38Cincludes transistors 2640.4 and 2642.4 (i.e., an n-type and a p-typetransistor, respectively) along with inverters 2644.4, 2646.4, and2648.4. Transistor 2640.4 receives a D input signal 2650.4 andtransistors 2640.4 and 2642.4 receive a clock (CLK bar) signal 2652.4,with inverters 2646.4 and 2648.4 providing a Q signal 2654.4 and a QN (QNOT) signal 2656.4, respectively.

FIG. 38D shows an exemplary circuit implementation for an XOR logic gate(e.g., XOR gate 2606.4 of FIG. 38 and FIG. 38A or an XOR gate 2717.4 asshown in FIG. 38D), which requires less power than a conventional XORgate. FIG. 38D includes inverters 2700.4 and 2702.4 and transistors2704.4 and 2706.4. An input signal (A) 2708.4 and an input signal (B)2710.4 are provided, as shown, to transistors 2704.4 and 2706.4,respectively, with inverter 2702.4 providing an output signal (Z) 2712.4in accordance with the equationZ=A·{overscore (B)}+Ā·B.

Likewise, FIG. 38E shows an exemplary circuit implementation for anexclusive NOR (XNOR) logic gate 2720.4, which may require less powerthan a conventional XNOR gate. FIG. 38E includes inverters 2722.4 and2724.4 and transistors 2726.4 and 2728.4. An input signal (A) 2730.4 andan input signal (B) 2732.4 are provided, as shown, to transistors 2726.4and 2728.4, respectively, with inverter 2724.4 providing an outputsignal (Z) 2734.4 in accordance with the equationZ={overscore (A·{overscore (B)})}+Ā·B.

Additionally, FIG. 38F shows an exemplary circuit implementation for aninverting multiplexer 2740.4, which includes transistors 2742.4 and2744.4 and an inverter 2746.4. Inverting multiplexer 2740.4 receivesinput signals (A) 2748.4, (B) 2750.4, and (C) 2752.4 and provides anoutput signal (Z) 2754.4, in accordance with the equationZ={overscore (C·B+{overscore (C)})}·A.

FIG. 38G shows an exemplary circuit implementation for a binary to Greyconversion (BGC) 2770.4 and a Grey to binary conversion (GBC) 2780.4.BGC 2770.4 shows an exemplary 4-bit conversion from binary to Grey codeutilizing XOR gates 2772.4, while GBC 2780.4 shows an exemplary 4-bitconversion from Grey to binary code utilizing XOR gates 2772.4. XORgates 2772.4 may be implemented as discussed above in reference to FIG.38D to minimize the amount of power utilized.

Another alternative implementation, in accordance with an embodiment ofthe present invention, for performing pulse width modulation includes alow frequency digital phase locked loop (DPLL) voltage controlledoscillator (VCO) with inverter stages and taps (i.e., example 1 aslisted above). FIG. 38H illustrates an exemplary implementation inaccordance with this embodiment, which includes a phase frequencydetector (PFD or phase detector) 2802.4, a charge pump 2804.4, a loopfilter 2806.4, a VCO 2808.4, and a frequency divider 2810.4.

PFD 2802.4 receives a reference frequency (e.g., 32 kHz) on a line2814.4 and a feedback signal from frequency divider 2810.4 on a line2812.4 and determines whether to provide a pump up or a pump down signalto charge pump 2804.4. Charge pump 2804.4 provides a signal, based onthe pump up or the pump down signal to loop filter 2806.4 and VCO2808.4. VCO 2808.4 is comprised of a ring oscillator having inverterstages and taps, as discussed similarly above in reference to FIG. 3,and therefore the description will not be repeated. An output signalfrom VCO 2808.4 is received by frequency divider 2810.4, which dividesdown a frequency of the output signal to a desired frequency to provideas the feedback signal.

Another alternative implementation for performing pulse width modulationmay be viewed as a hybrid of a high frequency DPLL with a countercombined with a digital comparator (i.e., example 2 as listed above) anda DPLL combined with a content addressable memory (i.e., example 3 aslisted above) to generate the required pulse width modulated signals.

More specifically, FIG. 38I illustrates a circuit 2900.4 to assist ingenerating pulse width modulation signals. Circuit 2900.4 includes adecoder 2902.4 which controls logic gates 2904.4 for writing to memory2906.4 (e.g., random access memory) having address lines (ADR) and datalines (D1). Memory 2906.4 stores data that determines start and stoptimes for desired pulse width modulated signals.

Memory 2906.4 provides data (at a DO terminal) under control of acontroller 2924.4 (through a read address (RA)) to multiplexers 2912.4,exclusive OR (XOR) gates 2910.4, and multiplexers 2918.4 viamultiplexers 2908.4. Under direction of controller 2924.4, XOR gates2910.4, multiplexers 2912.4 and 2918.4, registers 2914.4 and 2920.4, andadder/subtractor 2916.4 determine the data to provide to a comparator2922.4 to compare with a reference count 2926.4. Comparator 2922.4,which is similar to comparator 2310.4 of FIG. 36A, provides an outputsignal to a pulse width modulation circuit (i.e., PWM circuit 2314.4 asdescribed in reference to FIG. 36A). Consequently, by using memory and acomparator, pulse width modulated signals can be generated.

Section 1.1.6 Discussion of Phase Offset Tuning of Synchronous Drive toOptimize Conduction Losses vs. Dead Time

One of the efficiency optimizations in switching power supplies is toattempt to minimize the power dissipated by the Schottky diode that willtypically either be inserted in the circuit across the lower transistor(QB₁ in circuit 1301.2 of FIG. 46), or as is well known is intrinsic toa FET. In FIG. 46, the Schottky diode is indicated by referencecharacter S1. In order to minimize the voltage drop across diode S1,transistor QB₁ is inserted between terminal S and ground, and is turnedon in some instances to reduce the voltage drop across diode S1,therefore substantially reducing the power dissipated by this diode. Ofcourse, any of this power dissipated will not be delivered to the loadand so is therefore a source of inefficiency. The challenge becomesminimizing the amount of time that diode S1 conducts. It is desirablethat diode S1 conduct only a very small amount of time, however, if theswitching signals to QB₁ and QT₁, become too close in time, because ofthe time delay required to switch on and off, it's possible that bothtransistors might be on simultaneously which would not only be a hugeefficiency loss, but could potentially result in catastrophic circuitfailure due to the unlimited currents flowing directly from the inputsupply through QT₁ to QB₁ to ground. So the optimization then is to movethe gate signal for QB₁ as close as possible to the gate signal of QT₁,but without causing an overlap in conduction.

FIG. 39 illustrates the expected voltage at terminal S in FIG. 46 forthe switching timing diagrams immediately above it in the figures. Itwill be appreciated that the voltage at terminal S falls as the gatedrive to QT₁ is turned off and drops to the point where it is caught bythe Schottky diode S1, then in the subsequent time QB₁ is turned onpulling the voltage back up to the supply rail. This process happens inreverse order at the other end of the pulse where the gate for QB₁, isturned off. Current is again allowed to flow into diode S1, and thevoltage at terminal S falls below ground which results in wasted powerdissipated until QT₁ is turned on and the voltage at terminal S switchesup to the positive supply rail and the cycle repeats.

In prior art solutions, a guard band must be constructed between thetiming of these two gate signals. This guard band has to be long enoughto accommodate the slowest transistor that might be used in theapplication. In the present invention, in one embodiment we use dataprovided by the designer of the power supply being regulated byswitching power supply controller 1200 and program that data intointernal memory of switching power supply controller 1200, stating whatthe characteristics of the external transistors are. Accordingly, afixed guard band and time are not required. In a first embodiment, weuse the minimum number that could be used based on the published datafor the individual devices, and store the phase offset values in REGhw.In another embodiment of the present invention, we dynamically optimizethis further. This can be done by observing the efficiency of the givenoutput stage. For example, we know that for a buck power supply, theoutput voltage is equal to the duty cycle multiplied by the input supplyvoltage. Any deviation of the actual output voltage from that calculatedoutput voltage will be due to parasitic effects of inductors, resistors,capacitors, and the transistors themselves. It is therefore possible tocarefully, and at a slow rate over many cycles, tune the timing betweenQT₁ and QB₁ until this efficiency peaks and begins to degrade. It willdegrade rapidly as the transistors overlap, but this permits dynamicadaptation of the timing of the gate drive signals to the transistors tothe actual devices in the circuit being regulated. Switching powersupply controller 1200 can have a number of guard bands on a circuit tocircuit basis because the switching times of these transistors are alsosomewhat dependent on the load current that they are supplying and thetemperatures at which they are operating. The required amount of phaseoffset between the time QT₁ turns off and QB₁ turns on and the phaseoffset between when QB₁ turns off and QT₁ turns on may be different.Accordingly, fine tuning on both can be made. Temperature data isavailable to the controller from the Kelvin temperature sensor locatedin internal voltage supply 1209. In the first embodiment describedabove, providing the rise and fall time parameters for the transistorsto the controller can achieve a significant improvement over the priorart worst case tolerance scheme which does not take into account thespecifications of the actual transistors being used. In the secondembodiment, efficiency can be further improved by dynamically adaptingto the devices that are used, and thus achieving the highest efficiencypossible for the given set of transistors and discrete outputcomponents.

Section 1.1.7 Synchronous Sampling Multiple Output

Sample and Hold Module (SHM) 1207 samples various voltages and currentsand holds them until analog to digital converter 1206 is ready toreceive them. As noted herein, analog input signals used in the controlloops are provided by Digital Pulse Converter 1201 via SHM 1207, and areconverted to digital signals by analog to digital converter 1206. As anexample, SHM 1207 samples and holds thirteen voltages and sevencurrents, with seven voltages associated with external power suppliesbeing controlled or driven, two voltages associated with SUPPLYA andSUPPLYB and four auxiliary voltages: VOUT from Kelvin Temperature Sensor3516.4, and three pads TEMPEXT, AUX0 and AUX1.

FIG. 40 is a functional block diagram of SHM 1207 in one implementation.The functional block diagram includes an input/output (I/O) circuit3008.4, a voltage divider (scalar) 3010.4, a multiplexer 3012.4, and amultiplexer 3016.4. An external voltage is sampled by I/O circuit 3008.4via a pad 3000.4, with an output signal (OUTV) provided to voltagedivider 3010.4. Voltage divider 3010.4 divides down or scales the outputsignal (OUTV) to an acceptable level under control of a scale[3:0]signal received from regulation control module (REG) 1204. For example,an external voltage (e.g., 15 V or less) is scaled to an appropriatevalue for analog to digital converter 1206 (e.g., input range of 0 to3.3. V).

I/O circuit 3008.4 receives a SSUP signal and a SELVX signal fromregulation control module (REG) 1204 and an AUX signal and a SCLKVXsignal from DPC 1201. The AUX signal provides a sample clock forsampling one or more external supply voltages. The SSUP signal providesa select clock for selecting which external supply voltage to sample.The SCLKVX signal corresponds to SMPA[6:0] signal 2450.4 and SMPB[6:0]signal 2448.4, which control the sampling (and holding) of analogvoltages in SHM 1207 for analog to digital converter 1206. The SELVXsignal provides a select clock for selecting which external voltage tosample. An IDDQ signal provided to I/O circuits 3008.4 and 3016.4indicates a test control signal for these circuits.

An external current is sampled by I/O circuit 3016.4 via pads 3002.4 and3006.4 and utilizing a resistor 3004.4 connected between pads 3002.4 and3006.4, with an output signal (OUTC) provided. I/O circuit 3016.4receives an SWCAP[2:0] signal and an SELIX signal from regulationcontrol module (REG) 1204 and an SCLKIX signal from DPC 1201. TheSWCAP[2:0] provides input control for a switched capacitive network,discussed further below. The SCLKIX signal corresponds to SMPA[6:0]signal 2450.4 and SMPB[6:0] signal 2448.4, which control the sampling(and holding) of analog voltages (and currents) in SHM 1207 for analogto digital converter 1206. The SELIX signal provides a select clock forselecting which external current to sample.

Multiplexer 3012.4, under control of a MXSEL[4:0] signal from regulationcontrol module (REG) 1204, selects which input is presented to analog todigital converter 1206 as an output signal VINADC. Besides the inputsignals discussed above, multiplexer 3012.4 also receives a VOUT signalfrom a Kelvin temperature-sensor and a COUL0 and a COUL1 signals, whichare Coulombmetric measurements from available battery supplies (e.g.,battery zero and battery one, discussed further herein).

FIG. 40A is a functional schematic illustrating an exemplaryimplementation for I/O circuits 3008.4 and 3016.4. An external voltageor current is sampled via a pad 3030.4 and electrostatic dischargeprotection (ESD) circuitry 3032.4, with the sampled signal passingthrough a pass gate 3034.4 to a logic circuit 3036.4. Logic circuit3036.4 buffers or scales the sampled signal, for example, prior toproviding it to a capacitor 3038.4, which acts as a holding capacitoruntil a pass gate 3040.4 is opened and the sampled signal is provided toanalog to digital converter 1206 via a lead 3042.4 (labeled OUTV or OUTCfor output voltage or output signal from switched capacitor network).

In terms of general operation, the select clocks (i.e., the SCLKIX,SCLKVX, and AUX signals from DPC 1201) control pass gate 3034.4 forallowing an input voltage to charge capacitor 3038.4. The select lines(i.e., the SELIX, SELVX, and SSUP signals from regulation control module(REG) 1204) present the value stored on capacitor 3038.4 to be scaled(e.g., by voltage divider 3010.4) and passed to analog to digitalconverter 1206. If current is being measured, capacitor 3038.4 becomespart of a switched capacitor network to multiply a sampled input voltage(i.e., across resistor 3004.4) to a desired value for conversion byanalog to digital converter 1206.

FIG. 40B is a circuit schematic illustrating exemplary voltage andcurrent selection. An SHWIREI signal, an SHWIREV signal, and an SNHREFsignal is provided via pads 3060.4, 3062.4, and 3064.4 respectively. TheSHWIREI signal, SHWIREV signal, and SNHREF signal represent signals tobe measured by I/O circuits 3008.4 and 3016.4 (FIG. 40) forcorresponding current, voltage, and ground reference. As illustrated,the SCLKIX and SCLKVX signals control corresponding pass gates 3066.4,3072.4, 3076.4, and 3080.4 to allow an input voltage to charge acorresponding capacitor 3084.4 or 3086.4. The SELIX and SELVX signalscontrol corresponding pass gates 3068.4, 3074.4, 3078.4, and 3082.4 toallow the charge on corresponding capacitors 3084.4 or 3086.4 to pass asan output signal (labeled VOUTV or VOUTC for sampled voltage or current,respectively).

FIG. 40C is an exemplary interface signal block diagram for SHM 1207 inaccordance with another implementation. The interface signals include anIDDQ signal 3700.4, an AUX0 signal 3702.4, an AUX1 signal 3704.4, aTEMPEXT signal 3706.4, a VOUT signal 3708.4, a SUPPASENSE signal 3710.4,a SUPPBSENSE signal 3712.4, a SHWIREI[6:0] signal 3714.4, a SHWIREV[6:0]signal 3716.4, an SMPA[6:0] signal 3720.4, an SMPB[6:0] signal 3722.4,an SHNREF[9:0] signal 3724.4, a SELA[12:0] signal 3726.4, a SELB[6:0]signal 3728.4, a DIV[2:0] signal 3730.4, a SHM_CLK signal 3732.4, aMUXSEL[1:0] signal 3734.4, a DONE signal 3738.4, a VSEL_SMPA[8:0] signal3740.4, a ISEL_SMPA[6:0] signal 3742.4, a VREF_HALF signal 3748.4, aVSSIOA/B signal 3750.4, a VSS signal 3752.4, an AVS signal 3754.4, aVINADC signal 3758.4, an AVD signal 3760.4, a VDD signal 3762.4, and aVDDIOA/B signal 3764.4. Pads 3718.4 indicate signals received externalto switching power supply controller 1200.

IDDQ signal 3700.4 is a test signal, while AUX0 signal 3702.4 and AUX1signal 3704.4 are external signals for auxiliary applications. TEMPEXTsignal 3706.4 is received from a temperature sensor circuit attached tothe external battery. VOUT signal 3708.4 represents the voltage from theKelvin temperature sensor (e.g., Kelvin temperature sensor 2232.4 inFIG. 54) that will be sampled and held to isolate its ground from theground of analog to digital converter 1206 when this voltage getsconverted.

SUPPASENSE signal 3710.4 and SUPPBSENSE signal 3712.4 are used to sensethe voltage on the external Supply A and Supply B power supplies.SHWIREI[6:0] signal 3714.4 is used for sensing the voltage drop across asense resistor for each of the 7 external power supplies. SHWIREV[6:0]signal 3716.4 is used for sensing the voltage for each of the 7 externalpower supplies. SHNREF[9:0] signals 3724.4 are ground references forboth the voltage and the current of each external power supply, plusthere are three additional references for Supply A, Supply B, AUX0signal 3702.4, AUX1 signal 3704.4, and TEMPEXT signal 3706.4.

SMPA[6:0] signal 3720.4 is received from DPC 1201 and is used forsampling voltages at the load of the seven power supplies. SMPB [6:0]signal 3722.4 is received from DPC 1201 and is used for sampling thevoltage drop across the sense resistor for the seven supplies (describedin further detail below). SELA [12:0] signal 3726.4 is received fromregulation control module (REG) 1204 and determines which channelvoltage needs to be presented to a divider 3804.4 (FIG. 40D). SELB [6:0]signal 3728.4 is received from regulation control module (REG) 1204 anddetermines which channel current measurement.

VSEL_SMPA[8:0] signal 3740.4 is received from regulation control module(REG) 1204 and determines the swapping of the sample pulses for theleading edge. ISEL_SMPA[6:0] signal 3742.4 is received from regulationcontrol module (REG) 1204 and determines the swapping of the samplepulses for the falling edge.

DONE signal 3738.4 is received from analog to digital converter 1206when it is done with a conversion. DIV[2:0] signal 3730.4 is receivedfrom regulation control module (REG) 1204 and determines the dividervalue of divider 3804.4 for the voltage.

SHM_CLK signal 3732.4 is received from CLKGEN 1223, with a frequencythat is 16 times the frame frequency and is the value generated inCTS[4] within DPC 1201 as explained elsewhere herein. MUX_SEL [1:0]signal 3734.4 is used for selecting an output signal for a multiplexer3806.4 (FIG. 40D, e.g., an analog mux) that will be presented to analogto digital converter 1206. VDDIOA/B signals 3764.4 are the highestvoltages needed for substrate connection when using high voltageswitches. VSSIOA/B signals 3750.4 are the lowest voltages needed forsubstrate connection when using high voltage switches.

VDD signal 3762.4 (e.g., 3.3 V) is a digital voltage required for somecontrol logic. VSS signal 3752.4 is a digital ground required for somecontrol logic. AVD signal 3760.4 (e.g., 3.3 V) is an analog voltagerequired for some control logic. AVS signal 3754.4 is an analog groundto be tied to a ground of analog to digital converter 1206 whenconverting values into analog to digital converter 1206. VINADC signal3758.4 is provided to analog to digital converter 1206, for example,with a range between 0 and 3.0 V. VREF_HALF signal 3748.4 is a voltageoffset needed for a multiplier (discussed in detail below) to measureeither positive or negative voltage differences across analog to digitalconverter 1206. The value of VREF_HALF signal 3748.4 is one-half that ofVREF signal 3440.4 (FIG. 54A) and is received from IVS 1209. Theinterface signals are summarized in Table 1.1.7a.

TABLE 1.1.7a I/F Signal Type Description Source Destination IDDQ InputTest input if ITS [#] needed AUX0 Input I/O PAD Chip I/O auxiliarysample AUX1 Input I/O PAD Chip I/O auxiliary sample TEMPEXT Input I/OPAD Chip I/O external temp VOUT Input I/O PAD Chip I/O internal tempSUPPASENSE Input I/O PAD Chip I/O Supply A SUPPBSENSE Input I/O PAD ChipI/O Supply B SHWIREI [6:0] Input I/O PAD for Chip I/O current sampleSHWIREV Input I/O PAD for Chip I/O [6:0] voltage sample SMPA [6:0] InputSample pulse for DPC 1201 leading edge SMPB [6:0] Input Sample pulse forDPC 1201 falling edge SHNREF [9:0] Input I/O PAD ground Chip I/Oreference SELA [12:0] Input Select leading REG 1204 edge (voltages) SELB[6:0] Input Select falling RED 1204 edge (current) DIV [2:0] InputSelect divider REG 1204 value SHM_CLK Input CLK for sampling CLKGEN [#]MUXSEL [1:0] Input Analog Mux REG 1204 Select DONE Input ADC conversionADC 1206 finished VSEL_SMPA Input Selects leading REG 1204 [8:0] edgeswap ISEL_SMPA Input Selects falling REG 1204 [6:0] edge swap VDDIOAInput I/O Power for IVS 1209 HV transistors VDDIOB Input I/O Power forIVS 1209 HV transistors VSSIOA Input I/O Ground for IVS 1209 HVtransistors VSSIOB Input I/O Ground for IVS 1209 HV transistors AVSInput Analog Ground IVS 1209 AVD Input 3.3 V Analog IVS 1209 PowerVINADC Out- Data for the ADC ADC 1206 put VDD Input Global Digital 3.3 VVSS Input Global Ground 3.3 V VREF_HALF Input 1/2 VREF for the IVS 1209Multiplier

FIG. 40D is a functional block diagram 3788.4 of SHM 1207 in anotherimplementation. FIG. 40D is similar to FIG. 40 and therefore adiscussion regarding general operation will not be repeated. Blockdiagram 3788.4 of FIG. 40D includes an I/O circuit 3800.4, a multiplexer3802.4, a divider 3804.4, a multiplexer 3806.4, an I/O circuit 3810.4,and multiplexers 3808.4, 3820.4, and 3822.4.

The general function of block diagram 3788.4 is to sample nine voltages,seven of them being the external output voltage supplies and two of thembeing the Supply A and the Supply B, and also to sample the voltagedifference across a sense resistor (represented by a sense resistor3814.4 in FIG. 40D) for measuring current in each of the seven“regulated” output voltage supplies. Additionally, there are a fewsignals that are sampled on an as needed basis, for example, the voltageentering switching power supply controller 1200 as a result of atemperature sensor system that is connected to the battery (i.e., thevoltage of VOUT SIGNAL 3708.4), an internal voltage coming from Kelvintemperature sensor 2232.4 (FIG. 54) and the voltages of TEMPEXT signal3706.4, AUX0 signal 3702.4 and AUX1 signal 3704.4.

In general, block diagram 3788.4 interfaces with DPC 1201, regulationcontrol module (REG) 1204, and analog to digital converter 1206 andreceives from external sources three analog inputs (via pads 3818.4,3816.4, and 3812.4) that can be up to 15 V (e.g., for PDAs) or 16 V(e.g., for digital cameras). As shown in FIG. 40D, I/O circuits 3800.4and 3810.4 are coupled together to sample both voltage and current atthe same time, and convert these measurements at different times asneeded.

Multiplexer 3820.4 under control of ISEL_SMPA[6:0] signal 3742.4 selectsa signal from SMPA[6:0] signal 3720.4 and SMPB[6:0] signal 3722.4 andgenerates an SMPI signal 3821.4 for I/O circuit 3810.4. Multiplexer3822.4 under control of VSEL_SMPA[8:0] signal 3740.4 selects a signalfrom SMPA[6:0] signal 3720.4 and SMPB[6:0] signal 3722.4 and generatesan SMPV signal 3823.4 for I/O circuit 3800.4.

FIG. 40E illustrates an exemplary block diagram for I/O circuits 3800.4and 3810.4 in one implementation, which includes a number oftransmission gates 3824.4 and 3830.4, ESD protection 3825.4.4, one ormore capacitors 3828.4, along with a logic circuit 3826.4 having a testmechanism (via IDDQ signal 3700.4). FIG. 40E is similar to FIG. 40A andtherefore the general operation will not be repeated.

As shown in FIG. 40E, SHWIREI signal 3716.4 represents a voltage signalto be measured across sense resistor 3814.4 (FIG. 40D). As an examplefor sampling a current, a switched capacitor network will be utilizedfor the multiplication to be done for each of the seven currentsexplicitly (discussed in further detail below). SHWIREV signal 3714.4represents a voltage signal to be measured, and SHNREF signal 3724.4represents a voltage signal for a ground reference. The Supply A and theSupply B, along with TEMPEXT signal 3706.4 and VOUT signal 3708.4, willbe measured using a combination of SHWIREV signal 3714.4 and SHNREFsignal 3724.4 to sample and measure the voltages.

FIG. 40F is an exemplary circuit schematic illustrating voltage andcurrent selection for another implementation and may be viewed as anexpanded view of a portion of FIG. 40E. In terms of general operation(in reference to FIG. 40D and 1.1.7 g), the three basic analog inputsignals is as follows. When there is a sample pulse for SMPV signal3823.4, a measured voltage between pads 3816.4 and 3818.4 is stored in aholding capacitor 3858.4 as the voltage measurement at the load.

When there is a sample pulse for SMPI signal 3821.4, a measured voltagebetween pads 3812.4 and 3816.4 is stored across capacitors 3856.4,arranged in a parallel fashion by appropriate switching of pass gates3840.4, which represents the voltage measured across sense resistor3814.4. Note that the voltage difference across sense resistor 3814.4can be positive or negative depending on the external power supplytopology.

When a SELA signal corresponding to SELA[12:0] signal 3726.4 isasserted, the voltage in holding capacitor 3858.4 gets transferred intovoltage divider 3804.4 as a VOUTV signal 3912.4. When there is a SELBsignal corresponding to SELB[6:0] signal 3728.4, capacitors 3856.4 arestacked serially, by appropriate switching of pass gates 3840.4, toperform voltage multiplication (e.g., multiplication of 4) and provide aVOUTC signal 3859.4. The reference for the voltage multiplication isdone with respect to VREF_HALF signal 3748.4.

Divider 3804.4 divides the voltage down to a range of 0 to 3 V so thatit is within desired limits and analog to digital converter 1206 canread the value. Divider circuitry for divider 3804.4 will beimplemented, for example, using capacitor ratios with switches, twonon-overlapping clocks, and control logic. The non-overlapping clocksdischarge the capacitance of the divider network and prepare the dividernetwork for the next voltage sample.

FIG. 40G is an exemplary clock generation circuit in accordance withanother implementation. The two non-overlapping clocks, a DCLKH signal3884.4 and a DCLKL signal 3886.4, will be implemented by employing twoRS latches 3876.4 and 3878.4 and control logic, such as with an AND gate3880.4 and an OR gate 3882.4. The clock generation circuit receives DONEsignal 3738.4, SHM_CLK signal 3732.4, and DIV[2:0] signal 3730.4, asshown, to produce DCLKH signal 3884.4 and DCLKL signal 3886.4. Table1.1.7b provides exemplary divide or scalar values for given inputvoltages to produce representative values from analog to digitalconverter 1206.

TABLE 1.1.7b Divide Values (Scalar Input Voltages integers) (Voltage)ADC Value 1 0.498-2.502 166-834 2 2.508-5.502 418-917 4  5.508-11.508459-959 6 8.520-17.91 640-995

As an example, the maximum input voltage stored in holding capacitor3856.4 is 16 V. Depending upon the type of capacitor used (e.g., PiPcapacitors can only withstand 13.5 V), two or more capacitors may bestacked in accordance with an implementation of divider 3804.4.

FIG. 40H is an exemplary voltage divider 3898.4 in accordance withanother implementation for voltage divider 3808.4. VOUTV signal 3912.4is a selected channel voltage and switches 3900.4, 3902.4, 3904.4,3908.4, 3910.4, and 3912.4 are controlled by the non-overlapping clocks(DCLKH signal 3884.4 and DCLKL signal 3886.4) as shown. For example,because there are four values for division (i.e., divide values in Table1.1.7b), the default will be a divide by 1 which will allow VOUTV signal3912.4 to pass through directly, with DIV[2:0] signal 3730.4 set tologic 0. Voltage division is accomplished by charge sharing acrosscapacitors. Different effective capacitor values can be selected bytransfer gates 3900.4, 3902.4 and 3904.4 and their associated capacitors3914.4, 3916.4 and 3918.4. The DIV[2:0] determines which transfer gatesare selected to achieve the divide ratios. Table 1.1.7c providesexemplary divide values based on DIV[2:0] signal 3730.4.

TABLE 1.1.7c Divide DIV[2] DIV[1] DIV[0] Value 0 0 0 1 0 0 1 2 0 1 0 3 01 1 4 1 0 0 3 1 0 1 4 1 1 0 5 1 1 1 6 (default)

I/O circuit 3810.4 (FIG. 40D) includes a switched capacitor network toprovide voltage multiplication for a measured voltage difference acrosssense resistor 3814.4 for a corresponding current measurement. Forexample, in a continuous mode of operation at 2A and using a 0.1 ohmresistor for sense resistor 3814.4, the voltage difference across senseresistor 3814.4 is 200 mV. Thus, voltage multiplication based on thedesired voltage parameters is desirable. Furthermore, because theexternal battery may be charging itself from another power supplyconfiguration, the current sense may be negative with respect to theground of analog to digital converter 1206 and the multiplier circuit,illustrated below in reference to FIG. 40I, has the capability ofreversing polarities to always present a positive voltage difference tothe ADC.

Table 1.1.7d provides exemplary current measurements for a 2A powersupply system with a 0.1 ohm resistance for sense resistor 3814.4 (whereD.C.S. and C.S. stand for Deep Cycle Skipping and Cycle Skipping,respectively).

TABLE 1.1.7d Current Mode Method Voltage 4X 300 uA D.C.S. Dv/Dt  30 uV 120 uV  3 mA C.S. Dv/Dt 300 uV  1.2 mV  30 mA Discontinuous Digital  3mV   12 mV  2 A Continuous Direct 200 mV  800 mV

FIG. 40I is an exemplary voltage multiplier 3928.4 in accordance withanother implementation for I/O circuit 3810.4. Voltage multiplier 3928.4includes pass gates 3930.4 and capacitors 3934.4. In terms of generaloperation, when a sampling pulse from SMPI signal 3821.4 is received,capacitors 3930.4 are arranged in parallel by pass gates 3930.4. When aSELB signal of SELB[6:0] signal 3728.4 is asserted, capacitors 3934.4are stacked in series via pass gates 3930.4 and connected between eitherpositive or negative with respect to VREF_HALF signal 3748.4. It shouldbe noted that no matter what the polarity is for the voltage drop, itwill not affect the measurements since the reference is not ground butan offset voltage provided by VREF_HALF signal 3748.4.

Because there is a 4× multiplication done at once for each currentmeasurement channel (i.e., for the different current measurements foreach power supply), there will be 7 multipliers, with one for eachexternal power supply. FIG. 40J illustrates an exemplary block diagramof I/O circuit 3810.4 with various interface signals shown.

Multiplexer 3806.4 (FIG. 40D) employs high voltage switches and careneeds to be taken so that divider 3804.4 does not present a high voltagevia multiplexer 3806.4 to analog to digital converter 1206. FIG. 40K isa multiplexer scheme in accordance with another implementation thatmitigates the threat of a high voltage being presented to analog todigital converter 1206. As shown, SHM_CLK signal 3732.4 is gated withDIV[2:0] signal 3730.4, by an AND gate 4002.4, to properly present aDIVOUT signal 4010.4 from divider 3898.4 or VOUTC signal 3859.4 toanalog to digital converter 1206 (not shown) via a multiplexer 4008.4and VINADC signal 3758.4.

Section 1.2 Discussion of the Voltage Regulator, Detailed Hardware,Operations and Optimizations

FIG. 11 shows an alternative structure for implementing the principlesof this invention. In FIG. 11, controller 111 detects the currentthrough resistor R and the voltage across load capacitor CL. The chargeon load capacitor CL is used to drive a load on lead 112. The controller111 produces an error signal proportional to the voltage differencebetween the voltage across capacitor CL and a reference voltage, shownas system ground, but which could be any other desired referencevoltage. The output signal from controller 111 is supplied to an A to Dconverter 113 having ten output bits. This allows 1,024 levels to beidentified and quantized. The ten output bits from the A to D converterare sent on 10-bit bus 114 to comparator 115 where these bits arecompared to a reference number driven from programmable reference 116.Reference 116 is programmed by the user to contain the desired referencevoltage to be held by load capacitor CL. The output signal fromcomparator 115 is a binary difference signal D0 through D9 which istransmitted on 10-bit bus 117 to phase select circuit 118. Phase selectcircuit 118 is of the type, for example, shown above in FIG. 6. Thesignals D0 through D9 from comparator 115 are decoded using a circuitryof a type shown, for example, in FIG. 10 and are used to generate asignal which controls a pass transistor which allows a selected outputsignal from an inverter in the ring oscillator to be applied to the redinput lead 119 a of exclusive OR gate 119. The green input lead 119 b,as described above, will typically be the output signal pulled from thefirst inverter in the ring oscillator string. The result is a pulsewidth modulated output signal on lead 119 c from exclusive OR gate 119,which is then used to charge capacitor CL through resistor R (FIG. 11).The current through resistor R is measured by a signal detected on leads111 a and 111 b and used in controller 111 to provide a measure of thecharge being provided to load capacitor CL.

Analog to digital converter 1206 (FIG. 12), which measures and digitizesthe voltage of an analog signal to 10-bits of resolution, can beimplemented by a combination of a custom low-power mixed-signal circuit(ADC) and a digital logic circuit for interfacing the ADC to externallogic circuits. Inputs and output signals to analog to digital converter1206 include both analog and digital signals. In one implementation, theanalog to digital conversion can operate at 10 Msps (i.e., Mega-samplesper second).

FIG. 14 is a block diagram showing interface signals of analog todigital converter 1206, according to one implementation. As shown inFIG. 14, analog to digital converter 1206 includes: (a) analogmeasurement interface (AMI) 1401, digital interface (DI) 1402, and powersupply interface 1403. Analog measurement interface 1401 includes analoginput signal 1401 a (VIN), which is sampled for analog to digitalconversion, and an analog voltage reference input signal 1401 b (VREF).

Digital interface 1402 comprises data output bus 1402 a (DOUT[9:0]),analog to digital conversion completion or “done” signal 1402 b (DONE),analog to digital conversion initiation plus reset signal 1402 c(START/RSTN), and clock signal 1402 d (CLK). In this implementation,data output bus 1402 a is a 10-bit bus which provides the result of theanalog to digital conversion. In this implementation, clock signal 1402d has a frequency of about 128 MHz and a duty cycle of about 50%. Whenthe asynchronous reset signal 1402 e is held at a low logic value,circuits in analog to digital converter 1206 are held in a reset state.Similarly, when power down signal 1402 f is held at a low logic value,the analog circuits in analog to digital converter 1206 are powereddown, and the digital logic circuits are placed in a low-power state.

Power supply interface 1403 includes analog power and ground references1403 a and 1403 b (AVD and AVS), and digital power and ground references1403 c and 1403 d (VDD and VSS), respectively. In one implementation,both the analog and the digital power references (i.e., AVD and VDD) areprovided at 3.3V±10%.

FIG. 15 is a block diagram showing the interface signals of kelvintemperature sensor (KTS) 1500, which is an absolute temperature sensorcircuit that produces an output voltage linearly related to theCIRCUIT'S absolute temperature. As shown in FIG. 15, Kelvin temperaturesensor 1500 receives analog power and ground reference voltages atterminals 1501 and 1502 and asynchronous power-down control signal (PDN)at terminal 1503, and provides output voltage VOUT at terminal 1504,which is linearly related to the temperature of the circuit between 0.0Vand 3.0V.

Section 1.2.1 Hardware Description Regulation Control Module (REG) 1204

Combined Section 1.2.1.1 Detailed Description of the REG Module andSection 1.2.1.2 Voltage/Current Feedback SPS—Hardware Portion

Referring now to FIG. 41 and FIG. 12, Regulation control module (REG)includes a microcontroller 500.1 configured to generate pulse widthmodulation (PWM) information for a plurality of switching powerconverters under the control of switching power supply controller 1200.This PWM information may include the switching times for power switcheswithin the PWM switching power converters as well as voltage and currentsampling times for each switching power converter. Regulation controlmodule (REG) provides the PWM and sample information to Digital Signalto Pulse Converter (DPC) 1201, which in turn generates signalscontrolling pulse rising and falling edges to implement the PWMinformation, as well as generating the sample pulses for voltage andcurrent sampling.

As described further herein, DPC 1201 may be implemented in a number ofways, e.g., it may be CAM-based, ring-oscillator-based,comparator-based, or RAM-based. The following description will assumethat DPC 1201 is in the CAM-based embodiment. However, it will beappreciated that regulation control module (REG) 1204 would operateanalogously should DPC 1201 have a non-CAM-based embodiment.

The frame rate implemented by DPC 1201 affects the required processingspeed of microcontroller 500.1. A DPC frame is illustrated in FIG. 37.For example, if seven switching power supplies are being controlled byswitching power supply controller 1200 and the DPC frame rate is 524KHz, the PWM information for each switching power converter may beupdated approximately every 2 microseconds, which is the correspondingDPC frame period. Accordingly, a regulation scheduler module 521.1 maydivide each DPC frame period into calculation periods corresponding tothe various switching power supplies being regulated so as to implementa pipelined timeslot-based approach to the calculations for the PWMupdates and voltage and current feedback conversion scheduling for eachswitching power supply under control. In addition, another calculationperiod may be required for each DPC frame for related calculations withrespect to “book-keeping” tasks, and the monitoring of the various powersupplies for the switching power supplies being regulated. Thus, shouldthere be seven switching power supplies under control at a DPC framerate of 524 KHz, there would be eight 250 ns timeslots (7 powersupplies+1 slot for bookkeeping). Thus, the microcontroller 500.1 wouldhave only 250 nanoseconds to calculate the PWM information (pulse width)for each of the seven switching power supplies. Regulation schedulermodule 521.1 receives switching power supply controller 1200 clocksignal 522.1 and CST signals 2458.4 and coordinates the timing of thevarious modules within regulation control module (REG) 1204 accordingly.If each calculation period may be completed in 32 arithmetic steps(corresponding to 32 cycles of internal clock 522.1), microcontroller500.1 would require an instruction execution rate of 128 MHz. Thearithmetic functions implemented in each instruction may include, forexample, any of add, subtract, multiply, magnitude comparison, andabsolute value. Microcontroller 500.1 may be implemented using any of anumber of architectures to achieve the required operation speed,including a RISC-based or a VLIW-based architecture (i.e., “reducedinstruction set computer”-based architecture or “very-long-instructionword”-based architecture). For example, in a VLIW-based embodiment,microcontroller 500.1 may include a VLIW computing engine 517.1 thatinterfaces with a decode ROM (i.e., read-only memory, not illustrated)for decoding the VLIW instructions. In addition, microcontroller 500.1may include a RAM 516.1 for storing data and parameter values. Registerfiles 518.1 store the regulation parameter values for the variousswitching power supplies under regulation and provides run-time registerresources for calculations taking place in VLIW engine 517.1. In oneembodiment, VLIW engine 517.1 includes a topology register which pointsto the beginning of a 32-instruction segment in memory (corresponding tothe number of instructions that can be executed within a time slot). Thetopology register is so-named as the regulation algorithm for eachtopology (e.g., buck, boost, SEPIC) can be stored at different32-instruction segments in memory. In addition, a 5-bit program countercan be provided for sequencing VLIW engine 517.1 through the32-instruction segment in memory.

Referring again to FIG. 37, for a given switching power converter,regulation control module (REG) 1204 computes the timings for the risingand falling edges for a PFET pulse 2410.4 and an SFET pulse 2412.4. Therising edge and falling edge of PFET pulse 2410.4 control the on and offtimes, respectively, of a given switching power converter's primary FETswitch as driven by NFET driver module 1202. Similarly, the rising edgeand falling edge of SFET pulse 2412.4 control the on and off times,respectively, of the given switching power converter's secondary FETswitch as driven by NFET driver module 1202. The rising edge and fallingedge of SMPA pulse 2414.4 may control the voltage sampling period usedby sample and hold module (SHM) 1207 to obtain voltage feedbackinformation from the corresponding switching power converter. Similarly,the rising edge and falling edge of SMPB pulse 2416.4 may control thecurrent sampling period used by SHM 1207 to obtain current feedbackinformation from the corresponding switching power converter. However,either pulse SMPA 2414.4 or SMPB 2416.4 may be used for voltage orcurrent feedback purposes.

The four pulses illustrated in FIG. 37 thus correspond to eightindependent rising and falling edge times. For example, the falling edgeof SMPA pulse 2414.4 may be programmed to coincide with the rising edgeof PFET pulse 2410.4 (PFTS), because SMPA pulse 2414.4 can be requiredto be completed no less than 150 nano-seconds prior to the rising edgeof PFET 2410.4. (This relationship allows sufficient time formicroprocessor 500.1 to calculate the required duration of PFET pulse2410.4 using the data sampled by SHM 1207 using SMPA pulse 2414.4 andconverted by analog to digital converter 1206.) Similarly, the fallingedge of SMPB pulse 2416.4 may be programmed to coincide with the fallingedge of PFET pulse 2410.4 (PFTR), thereby allowing sampling of the peakcurrent in external inductor that occurs immediately prior the fallingedge of PFET pulse 2410.4. The widths of sampling pulses (i.e., SMPApulse 2414.4 and 2416.4) can be programmed, for example, at 2 or 4nanoseconds. Because at least 50 nanoseconds are to be provided betweensampling a pulse and a transition in the PFET and SFET pulses, theeffective duty cycle range computed by microprocessor 500.1 is 10% to90%. Duty cycles of 0% or 100% can be achieved in the DPC by masking theedge transitions of the PFET or SFET pulses, as needed.

Regardless of the number of independent pulse edges regulation controlmodule (REG) 1204 must control, other objectives besides powerregulation may affect the scheduling of the various pulse rising andfalling edges for each controlled switching power converter within a DPCframe. For example, having two switching power supplies switchsimultaneously may generate electromagnetic interference (EMI) or otherundesirable effects. In addition, simultaneous switching output (SSO)requirements may limit the number of FETS that may be switched at anygiven time. Accordingly, regulation control module (REG) 1204 mayschedule the various pulse rising and falling edges to satisfy suchobjectives using an edge scheduler 510.1. As described above,microcontroller 500.1 may implement a pipelined regulation approach suchthat each DPC frame is divided into calculation periods to calculatepulse width information 505.1 for each switching power supply controlledby switching power supply controller 1200. In a given DPC frame, thenecessary calculations to update PWM and schedule the digitization ofthe related analog voltage and current feedback signals are performedwith respect to the preceding DPC frame. If SHM clock signal 3732.4 hasa frequency {fraction (1/16)}^(th) that of SPS clock 522.1, two feedbacksignal analog-to-digital conversions may be accomplished per calculationperiod. Thus, a given DPC frame may be divided as follows:

Slot 0 Slot 1 Slot 2 Slot 3 Slot 4 Slot 5 Slot 6 Slot 7 IPC CALC SUP0CALC SUP1 CALC SUP2 CALC SUP3 CALC SUP4 CALC SUP5 CALC SUP6 CONV SUP0CONV SUP1 CONV SUP2 CONV SUP3 CONV SUP4 CONV SUP5 CONV SUP6 CONV SUPACONV SUPBThus, during slot (or calculation period) 0, the conversions for thevoltage and current feedback signals for the zeroth switching powersupply occur. Then, VLIW engine 517.1 may perform the necessary PWMupdates to a register file 518.1 using the converted feedback signalsfor the zeroth switching power supply during calculation period 1. Then,DPC I/F 590.1 may perform the necessary PWM update using the convertedfeedback signals from the zeroth slot of the previous DPC frame. Inaddition, the conversions for the first switching power supply feedbacksignals may occur in calculation period 1. Then, VLIW engine 517.1 mayperform the necessary PWM updates to a register file 518.1 using theconverted feedback signals in the second calculation period, the DPC I/Fmay perform the PWM updates using the converted feedback signals fromthe first time slot of the previous DPC frame, and so on for theremaining switching power supplies. Note that in calculation period 7,an analog-to-digital conversion is made for the voltages of powerssupplies A and B. No corresponding calculation period need be scheduled(switching power supply controller 1200 does not regulate these supplyvoltages). Because these conversions occur in the seventh timeslot, thezeroth timeslot of the next DPC frame may be used by regulationscheduler 521.1 to schedule any required inter-process communication andbookkeeping. A separate state machine may be used to implement edgescheduler 510.1. Having received the pulse width information and samplecommands from microcontroller 500.1, edge scheduler 510.1 may thenschedule the various pulse rising and falling edges. For example,referring again to FIG. 37, each DPC frame may be divided into aplurality of counts, e.g., 1024 counts (from 0 to 1023). The timebetween these counts determines the maximum PWM resolution that may beimplemented. The various pulse edges may be scheduled according to thecounts a DPC frame is divided into. Thus, edge scheduling would compriseassigning each edge to a DPC frame count. A number of algorithms may beused to provide the appropriate edge scheduling. For example, should apulse edge from two switching power converters be scheduled for the samecount, edge scheduler 510.1 could delay one of the conflicting pulseedges by one or more counts. Sample pulses SPAS 2414.4 and SPBS 2416.4may be scheduled to have a rising edge one or more counts before therising edge and falling edges of PFET pulse 2410.4, respectively. Aspecial cycle skipping mode is enabled by the edge scheduler through aprogrammable register. This provides the ability to skip the generationof a PFET pulse during a specified cycle. This is accomplished bywriting the same value to DPC 1201 for rising and falling edges of PWMpulses thereby generating a 0% DC output for those pulses.

To minimize the need for edge scheduling, central processing module(SYS) 1205 initializes regulation control module (REG) 1204 withappropriate rising and falling edge schedules prior to normal operation.For example, central processing module (SYS) 1205 stores in memory thedesired voltage levels provided by the various switching powerconverters, the expected power drawn by the loads, and the associatedoperating characteristics such as inductances and capacitances used inthe switching power converters. From this information, centralprocessing module (SYS) 1205 may calculate an expected pulse width foreach switching power converter. In turn, central processing module (SYS)1205 may then assign pulse edges corresponding to these pulse widthsdistributed across a DPC frame to help reduce the need for edgescheduling. For example, assuming a pipelined approach has beenimplemented, the rising edge of PFET pulse 2410.4 for each switchingpower supply has been calculated by central processing module (SYS)1205. Central processing module (SYS) 1205 then provides these initialedge schedules and other information to regulation control module (REG)1204 through an internal bus 520.1, which in turn may comprise buses forSFR-mapped data as well as buses for memory-mapped data. A bus interface525.1 may be used to coordinate the flow of data between regulationcontrol module (REG) 1204 and central processing module (SYS) 1205 oninternal bus 520.1 and provide the coordinated data to microcontroller500.1. For example, internal bus 520.1 may comprise an SFR enable signalREG_SFR_EN 700.1 to indicate an SFR transaction is set to occur; an SFRaddress signal SFR_ADDR[7:0] 701.1; an SFR data out signalSFR_DATA_OUT[7:0] 702.1 from central processing module (SYS) 1205; anSFR write enable SFR_WR 703.1; an SFR read enable SFR_RD 704.1; an SFRdata out signal SFR_DATA_IN[7:0] 705.1 from central processing module(SYS) 1205; a memory-mapped enable signal REG_MEM_EN 706.1 to indicate amemory-mapped transaction is set to occur; a memory-mapped addresssignal MEM_ADDR[15:0] 707.1; a memory-mapped data out signalMEM_DATA_OUT[7:0] 708.1 from central processing module (SYS) 1205; amemory-mapped data write enable signal MEM_WR_N 709.1; a memory-mappeddata read enable signal MEM_RD_N 710.1; a memory-mapped data out signalMEM_DATA_IN[7:0] 711.1 from regulation control module (REG) 1204; a modesignal PD[1:0] 712.1, a REG-generated interrupt signal REG_INT 713.1;and the central processing module clock signal SYS_CLK 714.1. Bycoordinating the flow of these signals on internal bus 520.1, interfacemodule 525.1 allows central processing module (SYS) 1205 to configureregulation control module (REG) 1204 and monitor its operation.

Referring again to FIG. 12, regulation control module (REG) 1204receives digitized feedback information (representing voltage andcurrent samples) from each switching power converter from analog todigital converter 1206 as signal DOUT[9:0] 715. The analog voltage andcurrent samples digitized by analog to digital converter 1206 areprovided by SHM module 1207 based upon the rising and falling edge timescorresponding to pulses SPAS 2414.4 and SPBS 2416.4. Referring now toFIG. 40D, a multiplexer 3806.4 within SHM 1207 may select between thesampled-and-held voltages and provide the selected voltage to analog todigital converter 1206. To keep analog to digital converter 1206operating efficiently, a conversion scheduler 540.1 receives the timinginformation from signals SMPA 2406.4 and SMPB 2408.4 and commandsmultiplexer 3806.4 accordingly using MUX_SEL[1:0] signal 3734.4. Asdiscussed above, if a pipelined approach is implemented, two conversionsmay be accomplished for each calculation period in a DPC frame.Conversion scheduler 540.1 drives SHM module 1207 to perform therequired scaling of the analog feedback signals and properly schedulethe digitization of the converted values in analog to digital converter1206.

Analog to digital converter 1206 may respond to a START/RSTN signal586.1 from conversion scheduler 540.1 toggling LOW to HIGH to begin theanalog-to-digital conversion process for a given analog feedback signal.During times of no active ADC activity, REG may pull START/RSTN signal586.1 LOW to put analog to digital converter 1206 into a low power mode.To keep the sampled voltages within the dynamic range of analog todigital converter 1206, conversion scheduler 540.1 provides a scalarvariable, DIV[1:0] 3730.4, to SHM 1207 to provide the appropriatescaling. Conversion scheduler 540.1 may drive SHM 1207 to select theappropriate voltage feedback signal for conversion through selectionsignal SELA[12:0] 3726.4. Similarly, selection signal SELB[8:0] 3728.4drives the appropriate selection of current feedback signals. Becauseeach pulse SMPA 2406.4 or SMPB 2408.4 may be used for either voltage orcurrent feedback, conversion scheduler 540.1 indicates which pulse hasbeen used for voltage or current feedback by signals VSEL_SMPA[8;0]3728.4 and ISEL_SMPA[6:0] 3742.2, respectively.

During a given DPC frame, multiplexer 3806.4 will begin to receive thevarious sampled-and-held voltage and current feedback signals.Conversion scheduler 540.1 may simply cause multiplexer 3806.4 to selectthe sampled-and-held feedback voltages as received in real time.Alternatively, conversion scheduler 540.1 may delay the schedulingaccording to whether or not all the feedback information has beenreceived from a given switching power converter. For example, supposethe voltage feedback from a given switching power converter has beensampled-and-held and received by multiplexer 3734.4. Conversionscheduler 540.1 would not cause multiplexer 3734.4 to select thisvoltage feedback signal, however, until the corresponding currentfeedback signal has also been received at multiplexer 3734.4.

To allow sufficient setup time for SHM 1207 to properly latch data,conversion scheduler 540.1 may switch the various SHM 1207 controlsignals described above one SPS clock cycle 522.1 before calculationperiod boundaries within a given DPC frame. In addition, conversionscheduler 540.1 may switch MUX_SEL[1:0] signal 3734.4 one-half SPS clockcycle 522.1 after values for SHM 1207 control signals SELA[12:0] 3726.4,SELB[6:0] 3728.4, and DIV[1:0] 3730.4 have been switched to protectanalog to digital converter 1206 from receiving any dangerous voltagefluctuations. In response to START signal 586.1, SHM module 1207 willgenerate the actual START pulse 3768.4, as shown in FIG. 40C, to analogto digital converter 1206 to begin the conversion. Analog to digitalconverter 1206 signals the completion of the conversion to regulationcontrol module (REG) 1204 using signal DONE 3738.4.

In addition to managing the scheduling of the feedback signals from theswitching power supplies, conversion scheduler 540.1 may also manage thesampling and conversion of the external supply A and B voltages, and thevoltages representing external and internal temperatures. Upon requestfrom central processing module (SYS) 1205, regulation control module(REG) 1204 schedules a conversion of the external and internaltemperature voltages (TEMPEXT signal 3706.4 and VOUT signal 3708.4,respectively, as illustrated in FIG. 1.1.7 d). When the conversion iscomplete regulation control module (REG) 1204 writes the data to astatus register which is read by central processing module (SYS) 1205 sothat the converted temperature values may be used in couloumbmetricmeasurements.

Conversion scheduler 540.1 controls the conversion of feedback samplesas well as source identification of the ADC data—i.e, whether or not itis a voltage or current sample and to which switching power supply itcorresponds as sample data DOUT[9:0] 715.1. However, in one embodiment,pulse widths are not adjusted if the voltage and current samples for agiven switching power supply are within a desired operating range. Thiscondition does not require VLIW engine 517.1 to do computations, thussaving power. Thus, regulation control module (REG) 1204 may includelimit comparison module 560.1 to test whether or not a given feedbacksignal is within the desired operating range (which may be denoted asthe deadband limits). If DOUT[9:0] 715.1 is within limits, conversionscheduler 540.1 may indicate this condition to microcontroller 500.1 sothat no PWM adjustment calculations need be performed for the associatedswitching power converter, and no DPC 1201 writes are required. Inaddition, limit comparison module 560.1 may also test whether DOUT[9:0]715.1 is within acceptable high and low regulation limits. If DOUT[9:0]715.1 is within these limits, limit comparison module 560.1 directs VLIWengine 517.1 to calculate new PWM values as described above, and valuescalculated for DOUT[9:0] 715.1 being within limits, are written to DPC1201. If DOUT[9:0] 715.1 is outside these limits, limit comparisonmodule 560.1 directs VLIW engine 517.1 to calculate new PWM values asdescribed above, and values calculated for DOUT[9:0] 715.1 being outsidelimits, are written to DPC 1201.

In response to assertion of DONE signal 3738.4, limit comparison modulelatches in DOUT[9:0] 715.1. Although conversion scheduler 540.1 andlimit comparison module 560.1 are illustrated as functionally separatefrom microcontroller 500.1, these functions may be performed bymicrocontroller 500.1 or by separate state machines.

Having received sample data DOUT[9:0] 715.1 from limit conversion module560.1, microcontroller 500.1 performs the pulse width calculations asdescribed above. As explained in greater detail with respect to DPC1201, sample scheduler 510.1 schedules the corresponding pulse edgesthrough data words DWI[19:0] 2428.4, their addresses ADW[4:0] 2430.4,and control signal WE 2432.4 that will be written into memory in DPC1201 through a DPC interface 590.1. DPC 1201 stored data may be read byregulation control module (REG) 1204 for testing or other purposes viaDRO[19:0] 2442.4, address ADW[4:0] 2430.4, and control signal RE 2434.4through DPC interface 590.1 As controlled by mode signal PD[1:0] 712.1from central processing module (SYS) 1205, regulation control module(REG) 1204 may be configured to implement a low-power mode as well asthe normal operation described herein. In this low-power mode,microcontroller 500.1 and edge scheduler 510.1 are powered down by,e.g., gating off the relevant clock signals such that regulation controlmodule (REG) 1204 controls only a subset of the possible switching powerconverters. Regulation control module (REG) 1204 subsequently passes onthe PD signal to DPC 1201 and CLKGEN 1223 as signal PD_OUT[9:0], whichmay be delayed from the time of receipt of PD[9:0] from centralprocessing module (SYS) 1205 via bus 520.1. A low power engine 585.1performs the necessary pulse width and sampling calculations. Forexample, low power engine 585.1 may issue a conversion request toconversion scheduler 540.1 to receive feedback data (voltage and/orcurrent) for a given switching power converter. To calculate therequired pulse widths based upon the received feedback information, lowpower engine 585.1 may perform a limit comparison as discussed withrespect to limit comparison module 560.1. If the sample is within adesired operating range, low power engine 585.1 may skip a number of DPCframes, e.g., four frames before again sampling feedback from theparticular switching power converter. If the sample is outside thedesired operating range, sleep engine will command DPC 1201 to changethe SFET 2404.4 and PFET 2402.4 pulse edges accordingly. To save power,the change may be based upon predetermined values stored in registers(not illustrated) associated with sleep engine 540.1 such that theimplemented PWM adjustment algorithm is a table look-up function.

In this low-power mode, CAM function in DPC 1201 is also disabled asdescribed further herein with respect to DPC 1201 operation. Thus, lowpower engine 585.1 directly causes the desired pulse edge times throughSET[28:0] and RESET[28:0] signals 2438.4 and 2440.4. Low power engine585.1 may be implemented through a configured state machine or othersuitable means.

x.2 LED Controller

Because switching power supply controller 1200 may be used inLED-containing devices such as PDAs, it may include LED control block,contained in central processing module (SYS) 1205 as shown in FIG. 12.FIG. 52 is a block diagram of LED control block 1214.

An interface module 30.1 receives LED control commands from centralprocessing module (SYS) 1205 (FIG. 12). In turn, interface module 30.1controls one or more LEDs using, e.g., first LED driver module 35.1 anda second LED driver module 40.1. Each driver module 35.1 and 40.1provides a PWM-modulated drive signal to its external LED (notillustrated).

The PWM modulation in each driver module 35.1 and 40.1 is under thecontrol of values written into registers within interface module 30.1.The greater the pulse width specified in each PWM period, the greaterthe amount of brightness shown by the corresponding LED. In addition,other registers within interface module 30.1 may control LED blinking,or ramp modes wherein a selected driver module gradually changes its PWMso that its corresponding LED gradually changes from one brightnesslevel to another.

X.4 Internal Power Supply Structure (GM)

Internal voltage supply (IVS) 1209 (FIG. 12) provides the operatingvoltages and power for internal operations in switching power supplycontroller 1200. IVS 1209 receives and provides various interfacesignals, including a reset (RSTn) signal, a power on reset (POR) signal,a clock output (CLK_OUT) signal, a clock input (CLK_IN) signal, a supplyA signal, a supply B signal, a battery 0 (batt0) signal, a battery 1(batt1) signal, and various supply or reference voltages (i.e., AVD,AVS, VDD, VSS, VDDIO[A,B], and VSSIO[A,B]).

The reset signal and the power on reset signal are received from resetmodule 1215 to reinitialize or reset operations of IVS 1209, inaccordance with one embodiment, or the power on reset signal may beprovided by IVS 1209 to other circuits within switching power supplycontroller 1200, in accordance with another embodiment. The clock inputsignal is received from external circuitry and a clock output signal isprovided to external circuitry to synchronize and enable variousclocking operations.

The supply signals (supplyA and supplyB) and the battery signals (batt0and batt1) are various external power supplies that may be received byIVS 1209. These external power supply sources, such as the exemplaryones listed below in Table x.4a, are coupled to switching power supplycontroller 1200 (and consequently to IVS 1209) and are used to power upand generate internal voltages (e.g., 3.3 V) and are used as powersupplies for various circuitry, such as the core logic and to supply ordrive external power NFETs (e.g., with up to 15 V).

TABLE x.4a Exemplary External Power Supplies EXTERNAL POWER SUPPLY MIN VMAX V Battery 0 or 1 2 Cell NiMH 1.8 3.8 4 Cell NiMH 3.6 7.6 Lion - 1Cell 2.7 4.2 Lion - 2 Cell 5.4 8.4 Supply A or B Car Adapter 9.6 14.4plus spikes Wall Adapter 4.5 15

IVS 1209 also includes a Kelvin temperature sensor (discussed below inreference to FIG. 54) to monitor the temperature of switching powersupply controller 1200 or external ICs. IVS 1209 also provides a voltagereference (VREF) that is used for various circuits, such as analog todigital converter 1206, and may also generate a power on reset signalfor distribution within switching power supply controller 1200, asexplained in further detail below.

FIG. 54 is an exemplary functional diagram of IVS 1209 in accordancewith one implementation. FIG. 54 shows the four external power supplies2202.4 through 2208.4 (i.e., BATT0, BATT1, SUPPLYA, and SUPPLYB,respectively) that are received via pads 2210.4 (with diodes 2212.4serving as protection devices, such as to guard against a reversepolarity condition). The BATT0, BATT1, SUPPLYA, and SUPPLYB labelsrepresent external battery source 0, external battery source 1, externalcar adapter or wall adapter source A, and external car adapter or walladapter source B, respectively.

IVS 1209 determines which of power supplies 2202.4 through 2208.4 touse, if more than one is available, to generate the core voltage and theI/O voltages for switching power supply controller 1200. For example,there may be two core voltages required (e.g., digital 3.3 V and analog3.3 V).

Switching power supply controller 1200 in accordance with oneimplementation has three modes of operation: 1) shut-down mode, 2)low-power mode, and 3) standard mode. In the shut-down mode, nothing onswitching power supply controller 1200 is powered on except whatmonitors the external sources of energy (e.g., power supplies 2202.4through 2208.4) and no current is flowing except leakage current inswitching power supply controller 1200.

In the low-power mode, internal and I/O voltages are maintained, DPCw1201 and two external loads (i.e., regulated power supplies) areoperating, and switching power supply controller 1200 is performingcycle skipping (discussed further herein). The amount of currentrequired is generally less than in the standard mode. The low-power modeis the default mode of operation once switching power supply controller1200 is powered-up from any other mode or when neither the clocks northe registers are running or set. In the standard mode, all of switchingpower supply controller 1200 is operating and all of the external loads(e.g., seven) are being supplied with regulated power. IVS 1209 receivecommands from central processing module (SYS) 1205 which will commandIVS 1209 to the appropriate mode.

As illustrated in FIG. 54, if power is being supplied by power supply2202.4 or 2204.4, then this external voltage is routed to and suppliespower to a crystal oscillator 2216.4, which generates a clock for avoltage multiplier 2214.4 (e.g., either a doubler or a tripler). Crystaloscillator 2216.4 has two pins, for input signals (XTALIN) and outputsignals (XTALOUT), via pads 2210.4.

An output of voltage multiplier 2214.4, which may be implemented usingwell known circuitry, is used as an input voltage for voltage regulators2226.4, 2228.4, and 2230.4, which supply voltages VDD, AVD, and VREFwhich can be of conventional circuitry. Additionally, voltage VREFH andVREFL are generated by circuits 2236 and 2238, respectively, andprovided to touch screen interface 1211. As will be appreciated from thefigures, to simplify the illustration, line have not been included toshow each connection. Instead, triangles are utilized within each blockindicating the signal and the direction of travel of the signal. Thesupply voltage VDD provides an output digital core voltage (e.g., 3.3 V)and is treated as global within switching power supply controller 1200.The supply voltage AVD provides an output analog core voltage (e.g., 3.3V) and is utilized by analog blocks that require a quieter power supplyvoltage than the supply voltage VDD. The supply voltage VREF shown onIVS 1209 and analog to digital converter 1206 in FIG. 12 also providesan output reference voltage for analog to digital converter 1206.Additionally, the supply voltages VDDIOA and VDDIOB are external I/Opower (A and B) used by internal NFET I/O buffers to drive the externalpower NFET. An external capacitor (labeled generically as EXT CAP) willbe generally connected to each of the corresponding bond pads (notshown). The supply voltage VDDIO3 is an external I/O power supply (e.g.,of 3.3 V) required for the general purpose I/O digital pads.

Once the supply voltages VDD and AVD are generated, a phase locked loop(PLL) 2218.4 operates and the supply voltages VDD and AVD aredistributed. A DONEMU signal is used to cause voltage multiplier 2214.4to cease its operations. A MODE[1:0] block 2224.4 informs voltageregulators 2226.4 and 2228.4 of the proper mode of operation and isrequired for setting the appropriate current.

The input voltage to voltage regulators 2226.4 and 2228.4 is alsoprovided to power on reset (POR) block 2234.4, for resetting orreleasing switching power supply controller 1200 via a RSTN_INT signal,and also to a charge pump block 2220.4 which may be implemented by anywell known charge pump circuit. Charge pump block 2220.4 stores theamount of charge (e.g., in external capacitors) required for aninstantaneous current needed to drive the external power NFET forsetting one or two regulated power supplies. A PUMPCLK signal and aDONEIO signal are control signals that cause charge pump block 2220.4when to start and when to stop operations, respectively.

If power is being received from power supply 2206.4 or 2208.4, crystaloscillator 2216.4 and voltage multiplier 2214.4 are bypassed and thesupply voltage is routed directly to voltage regulators 2226.4 through2230.4.

A band gap reference (BGR) block 2222.4 provides reference signals tovoltage regulators 2226.4 through 2230.4 and to a Kelvin temperaturesensor 2232.4. As noted above, voltage regulator 2230.4 generates thesupply voltage VREF. Kelvin temperature sensor 2232.4 generates a VOUTsignal.

FIG. 15 is a block diagram showing the interface signals of kelvintemperature sensor (KTS) 1500, which is an exemplary implementation ofKelvin temperature sensor 2232.4. Kelvin temperature sensor 1500 is anabsolute temperature sensor circuit that produces an output voltage(VOUT) linearly related to the circuit's absolute temperature. As shownin FIG. 15, Kelvin temperature sensor 1500 receives analog power andground reference voltages at terminals 1501 and 1502 and asynchronouspower-down control signal (PDN) at terminal 1503, and provides outputvoltage VOUT at terminal 1504, which is linearly related to thetemperature of the circuit between 0.0V and 3.0V.

FIG. 15A is an exemplary circuit for KTS 1500 in accordance with oneimplementation and highlights the basic equations that illustrate howthe output voltage VOUT corresponds to temperature. For example, all ofthe components in FIG. 15A can be integrated on the same chip and, thus,the matching of the components can be better than 0.1%. Withoutcalibration, approximately ±5% accuracy may be obtained over thetemperature range 0° C. to 125° C. With calibration, by measuring theoutput voltage VOUT at a known temperature (e.g., 25° C.), greater than±1% accuracy may be obtained.

FIG. 54A is an exemplary interface block diagram illustrating interfacesignals for IVS 1209 in accordance with another implementation and FIG.54B is a corresponding functional block diagram. As shown in FIG. 54Aand x.4c, the interface signals include a battery (BATT) signal 3400.4,a switch (SWT) signal 3402.4, power supplies A and B (SUPPLYA andSUPPLYB, respectively) 3404.4 and 3406.4, respectively, an XIN signal3408.4, an XOUT signal 3410.4, a VM1 signal 3412.4, a VM2 signal 3414.4,a VM3 signal 3416.4, an IVS_CLK signal 3418.4, a DONEPUMP signal 3420.4,a DONEBOOT signal 3422.4, a VOUT signal 3424.4, a RESETN signal 3426.4,a VSSIOA signal 3428.4, a VSSIOB signal 3430.4, a VDDIOA signal 3432.4,a VDDIOB signal 3434.4, a VREF_HALF signal 3436.4, a COLDBOOT signal, aSWT_ADAPT a FREF signal 3438.4, a VREF signal 3440.4, an AVS signal3442.4, a VSS signal 3444.4, an AVD signal 3446.4, and a VDD signal3448.4. Some of the signals employ pads 3550.4, as shown in FIG. 54A.

Battery (BATT) signal 3400.4 is the signal provided if an externalbattery is available. Switch (SWT) signal 3402.4 is used to causeswitching power supply controller 1200 to go from the shutdown mode intothe low power mode, or the standard mode. Power supply A 3404.4 andpower supply B 3406.4 are pads connected to the external power source (Aor B, respectively). XIN signal 3408.4 is an input signal for a crystaloscillator 3500.4, while XOUT signal 3410.4 is an output signal fromcrystal oscillator 3500.4, both for communication with an externalcrystal oscillator (e.g., of 32.768 kHz).

VM1 signal 3412.4, VM2 signal 3414.4, and VM3 signal 3416.4 are signalsprovided to and from external capacitors (labeled EXT CAP in FIG. 54B)coupled to a voltage multiplier 3504.4. IVS_CLK signal 3418.4 is a clocksignal from CLKGEN 1223 that is used by a charge pump 3518.4, whileDONEPUMP signal 3420.4 is a signal from central processing module (SYS)1205 to cause charge pump 3518.4 to stop. DONEBOOT signal 3422.4 is asignal from central processing module (SYS) 1205 to cause a powerdetection circuit 3502.4 to stop voltage multipler 3504.4 and voltageregulators 3510.4 and 3514.4.

VOUT signal 3424.4 is an output signal from a Kelvin temperature sensor3516.4, which is provided to SHM 1207. RESETN signal 3426.4 is a resetsignal from a power on reset (POR) circuit 3512.4 and is provided toreset module 1215. VSSIOA signal 3428.4 and VSSIOB signal 3430.4 are twoground signals, while VDDIOA signal 3432.4 and VDDIOB signal 3434.4 aretwo power supply signals generated in charge pump IO 3518.4, allprovided to various blocks in switching power supply controller 1200,including to NFET driver module 1202.

VREF signal 3440.4 is a reference voltage signal (e.g., 3.0 V) providedto analog to digital converter 1206, while VREF_HALF signal 3436.4 isone-half the voltage level of VREF signal 3440.4, which is provided toSHM 1207. FREF signal 3438.4 is a reference frequency provided bycrystal oscillator 3500.4. AVS signal 3442.4 is an analog ground signal,VSS signal 3444.4 is a digital ground signal, AVD signal 3446.4 is ananalog voltage signal (e.g., 3.3 V), and VDD signal 3448.4 is a digitalvoltage signal (e.g., 3.3 V).

As a functional example of operation (referring to FIG. 54B), if anexternal battery is supplying the power, battery signal 3400.4 willprovide a voltage (e.g., at least 1.8 V) to run power detection circuit3502.4 (which detects the voltage) and crystal Oscillator (XTAL OSC1)3500.4 to generate a clock for voltage multiplier 3504.4. Powerdetection circuit 3502.4 provides a voltage (e.g., 1.8 V) to crystaloscillator 3500.4 and detects when to exit the shutdown mode. The outputof voltage multiplier 3504.4 (e.g., twice the input voltage) is used asan input voltage for voltage regulators (VRAVD) 3514.4 and (VRVDD)3510.4, which generate a VRAVD voltage and a VRVDD voltage,respectively. A separate internal voltage doubler 3508.4 provides avoltage (e.g., 6.6 V or twice VDD signal 3448.4) to a VREF block 3506.4,which provides a stable reference voltage (i.e., VREF signal 3440.4) byemploying, for example, band gap reference circuits.

Once core logic VDD signal 3448.4 and AVD signal 3446.4 are generated, aphase-locked loop (PLL) circuit (not shown) along with other circuitryshown in FIG. 54B can begin operating using the voltage (e.g., 3.3 V)from VDD signal 3448.4 and/or AVD signal 3446.4. When the VRVDD voltagereaches a minimal level, a POR circuit 3512.4 exits a reset state(during reset, POR circuit 3512.4 functions to asynchronously set orreset registers used in switching power supply controller 1200). The8051 contained within central processing module (SYS) 1205 will thenbring up the rest of the system, which enables regulation control module(REG) 1204 to regulate an external supply (e.g., 3.3. V).

After regulation control module (REG) 1204 is regulating the externalvoltage supply (e.g., 3.3V), central processing module (SYS) 1205 willassert DONEBOOT signal 3422.4. Once DONEBOOT signal 3422.4 is asserted(e.g., logical high level), IVS 1209 will turn off voltage multiplier3504.4 and internal voltage regulators 3510.4 and 3514.4. The voltagefrom voltage regulator 3510.4 should also go to POR block 3512.4 so thatthe asynchronous registers (not shown) can exit their reset condition.

Additionally, charge pump 3518.4 will receive the output of voltagemultiplier 3504.4 and store in an external capacitor (EXT CAP) thecharge needed in NFET driver module 1202 during the power-up sequence(e.g., 3 V above the power supply to drive a gate of the external NFET).Charge pump 3518.4 will start once IVS_CLK signal 3418.4 is generatedand continue pumping until DONEPUMP signal 3420.4 from centralprocessing module (SYS) 1205.

If no battery is available and an external power source is provided bypower supply A 3404.4 or power supply B 3406.4 (e.g., either supplyproviding between 4.5 V and 14.4 V), this voltage will be detected byand run power detection circuit 3502.4 and also crystal oscillator3500.4. VREF block 3506.4 and voltage regulators 3510.4 and 3514.4 willreceive as their input voltage, either power supply A 3404.4 or powersupply B 3406.4 to generate the analog and digital voltages (i.e., AVDsignal 3446.4 and VDD signal 3448.4, respectively). Once AVD signal3446.4 and VDD signal 3448.4 are generated, the same procedure asdescribed above for a battery supply will take place to provide theexternal 3.3V connected back into the chip through AVD signal 3446.4 andVDD signal 3448.4.

If the chip has entered shutdown mode, and it still has power fromeither battery 3400.4, SUPPLYA 3404.4 or SUPPLYB 3406.4, it will remainin that state with just the Xtal Osc 3500.4 and Power Detection Circuit3502.4 running. It will remain in this state until one of two conditionsoccur: Pad SWT is pulled low or a supply is provided. Upon either orthese conditions will initiate the power up sequences described aboveand provide signal COLDBOOT and SWT_ADAPT to central processing module(SYS) 1205.

FIG. 54C is an exemplary flowchart for a power-up sequence for IVS 1209in accordance with another implementation. Step 3600.4 begins thepower-up sequence when an external voltage is applied. If the externalvoltage is greater than a required threshold (e.g., 1.8 V) at step3602.4, then the shutdown mode is held to allow circuits to power up(step 3604.4). If the external voltage is greater than expected batteryvoltages, then the external voltage is routed to VREF block 3506.4 andvoltage regulators (LDOs) 3510.4 and 3514.4 (as discussed in referenceto FIG. 54B).

If the external voltage is around the expected battery voltages, step3608.4 determines if a switch (SWT 3402.4) is pressed, if not then theshutdown mode (step 3604.4) is entered. If the power switch is pressed,step 3612.4 starts voltage multiplier 3504.4, VREF block 3506.4, andvoltage regulators 3510.4 and 3514.4. RESETN signal 3426.4 is asserted(step 3614.4) and charge pump 3518.4 operation begins.

Step 3618.4 determines if a low battery condition exists if so IVSreturns to step 3604.4. IVS remains in state 3618.4 until centralprocessing module (SYS) 1205 (referred to as 8051 in flowchart) hascompleted its coldboot operations. Once central processing module (SYS)1205 is done with these operations step 3620.4 switches off allredundant systems and the external voltage (i.e., VDD signal 3448.4) isfed back into switching power supply controller 1200. Step 3622.4 beginsoperation of voltage doubler 3508.4 and, when DONEBOOT signal 3422.4 isasserted, the standard mode or the low power mode (step 3626.4) isinitiated. When the shutdown mode command is received, step 3624.4switches off VREF block 3506.4 and voltage doubler 3508.4 and step3606.4 is repeated.

Section x.5 Converter Structure (ADC)

Referring to FIG. 12, analog to digital converter 1206 may beconstructed as a successive approximation converter (SAR) or in othersuitable architectures such a flash analog-to-digital converter. Due tothe use of sample and holds in combination with capacitor ratio basedvoltage division and multiplication; several unusual simplifications maybe made to the input of the analog-to-digital converter. A standardimplementation of a SAR converter requires an input buffer amplifier tocondition and scale the input voltage and a sample and hold circuit toprevent the input voltage from changing at the input to the SARcomparator as successive SAR DAC values are compared to the inputvoltage. In a SAR analog-to-digital converter, the conversion may becorrupted if the input voltage is allowed to change during theconversion process. The additional input stages normally required in aSAR analog-to-digital converter require that voltages to be measured bemade available and settled a significant time before the conversionstarts. This “set up time” is a particular issue in a multiplexed systemsuch as the present invention since it reduces the rate at which newinputs can be presented or demands much higher speed from theanalog-to-digital converter. Since the present invention utilizes sampleand holds at its inputs and capacitor ratio based scaling, both theinput amplifier and sample and hold sections may be eliminated from theSAR analog-to-digital converter implemented in an embodiment thuseliminating much of the set up time and several sources of error whilereducing the speed requirements and power consumption of the SARanalog-to-digital converter. The required processing speed of analog todigital converter 1206 is driven by the DPC frame rate and the number ofswitching power supplies under control by switching power supplycontroller 1200 and the desired A/D resolution. For example, ifswitching power supply controller 1200 uses a cycle rate of 524 KHz andcontrols seven switching power supplies, analog to digital converter1206 must convert 14 feedback signals plus two input signals in a DPCframe period of approximately 2 micro-seconds. If 10 bits of resolutionare desired, the resulting A/D logic clock frequency is approximately110 MHz.

x.7 Output Structure NFET (GM)

FIG. 55 is an exemplary functional diagram of NFET driver module 1202 inaccordance with one implementation. FIG. 55 includes input/output (I/O)buffers 2102.4 and 2104.4, which drive corresponding external powerMOSFETs 2112.4 and 2114.4 via pads 2106.4 and 2110.4 (corresponding toUPPER_FET and LOWER_FET terminals, respectively, on NFET driver module1202 of FIG. 12).

As shown in FIG. 55, power MOSFET 2112.4 is referred to as an “UPPERFET” and power MOSFET 2114.4 is referred to as a “LOWER FET,” with bothemployed to provide a regulated DC supply to a load. NFET driver module1202 receives via switch control bus 1303 a (i.e., HIGHFET) and switchcontrol bus 1303 b (i.e., LOWFET) signals shown adjacent to the bus,which contain information utilized by NFET driver module 1202 to controlI/O buffers 2102.4 and 2104.4. A pad 2108.4 is coupled to power MOSFETs2112.4 and 2114.4 to provide feedback (fb) to NFET driver module 1202regarding external load parameters (e.g., voltage).

NFET driver module 1202 (FIG. 12) further includes interface signalsVDDIO[A,B], VSSIO[A,B], UPFET_Source, LOWFET_Source, and IDDQ. TheVDDIO[A,B] signals are supply voltages A and B, while the VSSIO[A,B]signals are corresponding A and B ground references. The UPFET_Sourceand LOWFET_Source signals monitor and provide feedback regarding powerMOSFETs 2112.4 and 2114.4, respectively, and correspond symbolically tothe feedback (fb) illustrated in FIG. 55. The IDDQ signal is providedfor test purposes.

FIG. 55A illustrates exemplary interface signals of NFET driver module1202 in accordance with another implementation. The interface signalsinclude a PFET[6:0] signal 3200.4, an SFET[6:0] signal 3202.4, an IDDQsignal 3204.4, a SELMODE signal 3206.4, a CNTL[1:0] signal 3208.4, aVSSIOA signal 3210.4, a VSSIOB signal 3212.4, a SUPPLYB signal 3214.4, aSUPPLYA signal 3216.4, an UPSENSE[6:0] signal 3218.4, a LOWFET[6:0]signal 3220.4, an UPFET[6:0] signal 3222.4, a VDDIOB signal 3224.4, aVDDIOA signal 3226.4, a VSS signal 3228.4, and a VDD signal 3230.4.

PFET[6:0] signal 3200.4 and SFET[6:0] signal 3202.4 are the pulsesreceived from DPC 1201 to drive the primary and secondary externalNFETs, respectively. IDDQ signal 3204.4 is a test signal, while SELMODEsignal 3206.4 selects a mode of operation and CNTL[1:0] signal 3208.4provides control information.

VSSIOA signal 3210.4 and VSSIOB signal 3212.4 are return ground pathsfor the external power supplies (i.e., SUPPLYA signal 3216.4 and SUPPLYBsignal 3214.4, respectively). VDDIOA signal 3226.4 and VDDIOB signal3224.4 are high voltage sources derived from supply A 3216.4 or supply B3214.4, which may be controlled by DPC 1201 to regulate the load voltageas needed. VSS signal 3228.4 and VDD signal 3230.4 are digital groundand digital supply voltage (e.g., 3.3 V), respectively.

UPFET[6:0] signal 3222.4 and LOWFET[6:0] signal 3220.4 are signals thatdrive the primary and secondary external NFETs, while UPSENSE[6:0]signal 3218.4 is a reference source voltage for the primary externalNFETs of switching power converters being controlled by switching powersupply controller 1200.

FIG. 55B illustrates exemplary interface signals of switch driver modulealso referred to as NFET driver module 1202 in accordance with anotherimplementation. NFET driver module 1202 contains I/O drivers, asexplained below, that can be used to drive external power FETs as wellas an external coil (e.g., for digital camera applications). Theinterface signals include PFET[6:0] signal 3200.4, SFET[6:0] signal3202.4, a PFET_SEL[6:0] signal 4402.4, a FET_SWAP[6:0] signal 4404.4, anEN_EXT33 signal 4406.4, IDDQ signal 3204.4, a PFETSENSE[6:0] signal4408.4, a SUPPLY_SEL[6:0] signal 4410.4, VDDIOA signal 3226.4, VSSIOAsignal 3210.4, VDDIOB signal 3224.4, VSSIOB signal 3212.4, a PADIO[6:0]signal 4418.4, a PFETDRIVE[6:0] signal 4416.4, an SFETDRIVE[6:0] signal4414.4, and an EXTDRIVE signal 4412.4.

PFET[6:0] signal 3200.4 and SFET[6:0] signal 3202.4 are the pulsesreceived from DPC 1201 used to drive the external primary NFETs andexternal secondary NFETs, respectively. PFET_SEL[6:0] signal 4402.4,received from regulation control module (REG) 1204, commands whether todrive external FETs or external coils. If driving external coils, VDDIOAsignal 3226.4 or VDDIOB signal 3224.4 will be connected to the Supply Aor the Supply B. If driving external FETs, VDDIOA signal 3226.4 orVDDIOB signal 3224.4 will be boosted to voltage levels equivalent to 3 Vabove the voltage level of the Supply A or the Supply B.

FET_SWAP[6:0] signal 4404.4, received from regulation control module(REG) 1204, signals NFET driver module 1202 to swap the pulses fromPFET[6:0] signal 3200.4 and SFET[6:0] signal 3202.4, depending uponwhether a boost or a buck is the external power topology. EN_EXT33signal 4406.4 drives EXTDRIVE signal 4412.4, which drives the externalswitch to isolate the external voltage supply (e.g., 3.3 V) from therest of the system. IDDQ signal 3204.4 is a test signal.

PFETSENSE[6:0] signal 4408.4 is the source of the primary external NFETand is connected back into switching power supply controller 1200 tomonitor the source voltage of PFETDRIVE[6:0] signal 4416.4SUPPLY_SEL[6:0] signal 4410.4 selects between the Supply A and theSupply B for generation of VDDIOA signal 3226.4 or VDDIOB signal 3224.4.VDDIOA signal 3226.4 is connected externally to PADIO[6:0] signal 4418.4and provides a voltage level above the voltage of the Supply A (e.g.,3.0 V above) as a main power supply for PFETDRIVE[6:0] signal 4416.4 orSFETDRIVE[6:0] signal 4414.4 (if driving an external NFET) or as apre-driver voltage (if driving an external coil). VSSIOA signal 3210.4is a ground signal for VDDIOA signal 3226.4 or PADIO[6:0] signal 4418.4.Similarly, VDDIOB signal 3224.4 is connected externally to PADIO[6:0]signal 4418.4 and provides a voltage level above the voltage of theSupply B (e.g., 3.0 V above) as a main power supply for PFETDRIVE[6:0]signal 4416.4 or SFETDRIVE[6:0] signal 4414.4 (if driving an externalNFET) or as a pre-driver voltage (if driving an external coil).

VSSIOB signal 3212.4 is a ground signal for VDDIOB signal 3224.4 orPADIO[6:0] signal 4418.4. PADIO[6:0] signal 4418.4 is a main source ofinput/output power for the final stage in PFETDRIVE[6:0] signal 4416.4or SFETDRIVE[6:0] signal 4414.4 and may utilize VDDIOA signal 3226.4 orVDDIOB signal 3224.4 as well as the Supply A or the Supply B, dependingupon the configuration (discussed further below). PFETDRIVE[6:0] signal4416.4 drives the external primary FET. SFETDRIVE[6:0] signal 4414.4drives the external secondary FET. EXTDRIVE signal 4412.4 switches on oroff the external FET switch used for isolating the external power supply(e.g., 3.3 V) that will be connected back to switching power supplycontroller 1200.

FIG. 55C shows an exemplary functional diagram for the implementation ofFIG. 55B. In terms of general operation, two pulses, PFET signal 3200.4and SFET signal 3202.4 are received from DPC 1201. If these pulsesoverlap, a fail safe mode logic 4430.4 (FIG. 55D) sets the output signalto LOW to PFETDRIVE signal 4416.4 and SFETDRIVE signal 4414.4 until thenext frame. Pre-driver logic 4432.4 and 4434.4 (FIG. 55D) checks forswapping between external primary and secondary output terminals.PFET_SEL signal 4402.4 will also configure the driver either as drivingexternal FETs or driving directly the external coil. PFETDRIVE signal4416.4 needs to be, for example, 3 V higher than PFETSENSE signal 4408.4when driving external FETs. Hence, PFETSENSE signal 4408.4 is connectedback into the pre-driver logic to monitor the source voltage for theexternal FET.

VDDIOA signal 3226.4 and VDDIOB signal 3224.4 (FIG. X.7 d) are connectedexternal to NFET driver module 1202 and each of the channels (e.g.,seven) of NFET driver module 1202 will be connected to either VDDIOAsignal 3226.4 or VDDIOB signal 3224.4, depending on the value ofSUPPLY_SEL signal 3202.4. Each of the seven channels utilize PADIOsignal 4418.4, PFETDRIVE signal 4416.4, PFETSENSE signal 4408.4,SFETDRIVE signal 4414.4, and its corresponding VSSIOA signal 3210.4 orVSSIOB signal 3212.4. During the power-up sequence an external NFET willbe driven by EXTDRIVE signal 4412.4, which is an I/O buffer similar tothe buffers used for PFETDRIVE signal 4416.4. EN_EXT33 signal 4406.4,received from central processing module (SYS) 1205, controls EXTDRIVEsignal 4412.4.

FIG. 55D illustrates an application employing internal buffers,represented by transistors 4440.4 and 4442.4, to directly drive anexternal coil 4444.4 for the implementation of FIG. 55B. As discussedabove for this configuration, PADIO signal 4418.4 is connectedexternally (i.e., at the board level) to Supply A or the Supply B. Asshown, PFETDRIVE signal 4416.4 and SFETDRIVE signal 4414.4 may bearranged to drive in parallel external coil 4444.4 and a capacitive load4446.4, which may result in lower source impedance.

Fail safe mode logic 4430.4 verifies that PFET signal 3200.4 and SFETsignal 3202.4 do not overlap. Supply_SEL 4410.4 selects VDDIOA signal3226.4 or VDDIOB signal 3224.4 via a multiplexer 4448.4 to route tocontrol logic 4432.4 and 4434.4, which control transistor pairs 4440.4and 4442.4.

FIG. 55E illustrates an application employing internal buffers,represented by transistor pairs 4440.4 and 4442.4, to drive externalFETs 4450.4 and 4452.4 for the implementation of FIG. 55B. As discussedabove for this configuration, PADIO signal 4418.4 is connectedexternally (i.e., at the board level) with VDDIOA signal 3226.4 orVDDIOB signal 3224.4. The application shown in FIG. 55E is similar toFIG. 55D and therefore will not be repeated. However, as shown in FIG.55E, external FETs 4450.4 and 4452.4 are driven by PFETDRIVE signal4416.4 and SFETDRIVE signal 4414.4, respectively, to drive external coil4444.4 and capacitive load 4446.4. When PFET 3200.4 is high, its controllogic 4432.4 maintains PFETDRIVE at a constant 3V above PFETSENSE 4408.4

Table x.7a summarizes in a truth table format the state of varioussignals for the exemplary configurations. For PFET_SEL signal 4402.4, azero and a one indicate internal and external, respectively. ForFET_SWAP signal 4404.4, a zero and a one indicated no swap and swap,respectively.

TABLE x.7a PFET SFET PFET_SEL FET_SWAP PFETDRIVE SFETDRIVE 0 0 0 0 0 0(internal) (internal) 0 0 0 1 0 0 (internal) (internal) 0 0 1 0 0 0(external) (external) 0 0 1 1 0 0 (external) (external) 0 1 0 0 0 1(internal) (internal) 0 1 0 1 1 0 (internal) (internal) 0 1 1 0 0 1(external) (external) 0 1 1 1 1 0 (external) (external) 1 0 0 0 1 0(internal) (internal) 1 0 0 1 0 1 (internal) (internal) 1 0 1 0 1 0(external) (external) 1 0 1 1 0 1 (external) (external) 1 1 0 0 0 (fail0 (fail safe) safe) 1 1 0 1 0 (fail 0 (fail safe) safe) 1 1 1 0 0 (fail0 (fail safe) safe) 1 1 1 1 0 (fail 0 (fail safe) safe)

FIG. 55F shows an exemplary on-chip configuration block diagram for theimplementation of FIG. 55B. As shown, switching power supply controller1200 receives the Supply A or the Supply B signal via pads 4462.4 orthrough PADIO signal 4418.4, and generates VDDIOA signal 3226.4 andVDDIOB signal 3224.4. Switching power supply controller 1200, throughNFET driver module 1202, generates PFETDRIVE signal 4416.4 and SFETDRIVEsignal 4414.4 and receives PFETSENSE 4408.4 and optionally SFETSENSE4460.4.

x.8 Microcontroller Structure

Referring to FIG. 56, an exemplary embodiment for central processingmodule (SYS) 1205 comprises a microprocessor core 400.1 such as an 8051which interfaces with external hosts through an Advanced General PurposeI/O (AGPIO) module 410.1. ADGPIO module 410.1 provides 31 GPIO portsthat may be individually configured to suit a host application's needs,such as the ability to determine power supply status, an 8×8 keyboardinterface, and a serial communication bus for communication with a hostdevice.

Such serial communication with a host device may occur using a number ofsignaling protocols as selected by a serial multiplexer 420.1. Forexample, serial data may be received from a universal asynchronousreceiver/transmitter (UART) 425.1, an SPI receiver/transmitter 435.1, ora JTAG receiver/transmitter 440.1. Additional serial devices could alsobe used such as Microwire, I2C, or SSI2 devices. Microprocessor core400.1 communicates with the serial device selected by serial multiplexer420.1 on an internal bus 450.1.

A data-path multiplexer 470.1 multiplexes the data on a read data pathin internal bus 450.1 to select data from different sources such as LEDcontroller 1214 and watchdog controller 1213. In addition,microprocessor core 400.1 may receive data from and store data to thenon-volatile memory module 1216 (FIG. 12) through an external interface455.1 coupled to internal bus 450.1. As described further herein withrespect to watchdog controller 1213 and LED controller 1214,microcontroller 400.1 also interfaces with these modules throughinternal bus 450.1.

Data necessary for program execution on microprocessor core 400.1 may bestored in a program memory device 460.1. Program memory device 460.1 maybe any technology suitable for program storage and execution, such asmasked ROM, Flash memory, EEPROM, or other suitable media.Microprocessor core 400.1 may also store data in a RAM 465.1 or in anon-volatile memory module 1216 (FIG. 12). Microprocessor core 400.1communicates with other modules such as regulation control module (REG)1204 in switching power supply controller 1200 through internal bus520.1, which is also illustrated and discussed with respect toregulation control module (REG) 1204 in FIG. 12. A handshaking protocolor glue logic may be used to coordinate this communication. The outgoingdata is addressed to a particular module in a memory-mapped fashion orby SFR-mapped addresses.

Microprocessor core 400.1 is clocked by clock signal 714.1 received fromclock generation module 1223. However, because the regulation from DPCframe to DPC frame during normal operation is under the control ofregulation control module (REG) 1204, it would be a waste of power tohave microprocessor 400.1 clocked continuously by clock 714.1 duringnormal operation. Thus, clock 714 is gated off during normal operationby the microprocessor when the microprocessor has completed anyprocessing that is needed. In response to interrupts from other modulessuch as watchdog controller 1213 or LED controller 1214, the powermanagement module 480.1 allows microprocessor core 400.1 to receiveclock 714.1. Once the interrupt has been serviced, clock 714.1 is onceagain gated off by the microprocessor core 400.1.

Section 1.2.2 Control Loops/Algorithms

FIG. 25 is a diagram of switching power supply controller 1200 beingused in a battery and power supply management application in, forexample, a personal digital assistant (PDA). As shown in FIG. 25,switching power supply controller 1200 (a) regulates buck converter 2570to provide a regulated DC power supply at terminal 2540, (b) regulatesDC/AC converter 2571 to provide an AC power supply between terminals2542 and 2543, (c) to charge battery 2517 through converter 2572,operating either in buck or boost mode, when an external DC supplyvoltage (e.g., 12-15 volts) is available at terminal 2544, and (d) todraw power from battery 2517, operating in boost mode, when the externalDC supply voltage at terminal 2544 is not available. In a PDAapplication, for example, switching power supply controller 1200 maycommunicate with a host processor and peripheral devices over datainterface 2573. Conventional reference oscillator circuit 2574 providesa 32.768 KHz reference clock signal for switching power supplycontroller 1200.

Even though specific converter configurations are shown in FIG. 25 toillustrate the exemplary embodiments described herein, one skilled inthe art would appreciate that the present invention is not so limited.Other converter configurations can be implemented within the scope ofthe present invention based upon this detailed description. In FIG. 25,buck converter 2570 includes inductor 2503, current-sense resistor 2504,output capacitor 2501, and MOS switches 2505 and 2506 (which includesintrinsic diode 2502) receiving at their respective gate terminals thepulse-width modulated drive signals (“upper_FET gate” and “lower_FETgate”) from switching power supply controller 1200.

Inductor 2503, current-sense resistor 2504 and output capacitor 2501 areconnected in series between sense input terminal 2518 of switching powersupply controller 1200 (one of the “upper_FET_source_sense” terminals),which is connected to the source terminal of MOS switch 2505, and aground reference. Diode 2502 prevents the voltage at sense inputterminal 2518 from dropping below a predetermined voltage (e.g.,approximately 1 volt) less than the ground reference. The drain terminalof MOS switch 2505 is connected to one of two supply terminals (“SupplyA” or “Supply B”) of switching power supply controller 1200. The drainand source terminals of MOS switch 2506 are connected respectively tothe sense input terminal 2518 and the ground reference. The regulatedoutput voltage is taken from the ungrounded terminal 2540 of outputcapacitor 2501. This regulated output voltage is determined by the dutycycle of the pulse-width modulated signals received at the gateterminals of MOS switches 2505 and 2506. In one implementation, theperiod of the pulse-modulated signals (also referred to as a “cycle”) istwo microseconds, as shown in FIG. 22.

The terminals of current-sense resistor 2504 are connected to senseinput terminals 2530 and 2531 (one terminal from each of the “sense_I”and “sense_VI” buses) of switching power supply controller 1200. Thevoltage (V_(out)−V_(IL)) across these sense input terminals isproportional to the current in inductor 2503.

External DC power can be provided to supply terminal 2508 of the systemfrom terminal 2544. In converter 2572, diodes 2511, 2512 and 2513 areconfigured to ensure that power flows only from the external powersource into the system, and to prevent the voltage at supply terminal2508 from going below a predetermined voltage (i.e., the forward biasvoltage of the diode) less than the ground reference. In converter 2572,inductor 2514, sense-resistor 2515 and capacitor 2516 are connected inseries between sense terminal 2519 of switching power supply controller1200 (one of the “upper_FET_source_sense” terminals) and a groundreference. The ungrounded terminal of capacitor 2516 is connected to thepositive terminal of battery 2517. MOS switches 2509 and 2510 receivepulse-width modulated drive signals at their gate terminals. The drainand source terminals of MOS switch 2509 are connected respectively topower supply terminal 2508 and sense terminal 2519. The drain and sourceterminals of MOS switch 2510 are connected respectively to senseterminal 2519 and the ground reference. When an external power source isconnected at terminal 2544, the pulse-width modulated drive signals atthe gate terminals of MOS switches 2509 and 2510 regulate the voltageand current for charging battery 2517. Alternatively, i.e., when anexternal power source is not connected at terminal 2544 and the systemruns from power supplied by battery 2517, the pulse-width modulatedsignals at the gate terminals of MOS switches 2509 and 2510 regulate thevoltage at power supply terminal 2508. Depending on whether the voltagesupplied by battery 2517 is higher than the desired voltage at terminal2508, converter 2572 is operated as a buck or boost converter.

The terminals of current-sense resistor 2515 are connected to senseinput terminals 2535 and 2536 (one terminal from each of the “sense_I”and “sense_VI” buses) of switching power supply controller 1200. Thevoltage (V_(out)−V_(IL)) across these sense input terminals isproportional to the current in inductor 2514.

DC/AC converter 2571 provides a high voltage AC power supply (e.g., 700volts) for cold-cathode fluorescent lighting (CCFL) used asback-lighting in a PDA. In converter 2571, MOS switches 2521 and 2522alternatively connect inductors 2520 and 2523 to the ground reference.Inductors 2520 and 2523 are each connected between power supply terminal2508 and the drain terminal of one of MOS switches 2521 and 2522respectively. The voltages at these drain terminals controlpiezoelectric transformer 2524 to provide the requisite AC signal acrossterminals 2542 and 2543 of CCFL 2525. Terminal 2543 is connected to theground reference through sense-resistor 2526.

Section 1.2.2.1 Input Voltage/Output Current Predictive Control Loop

The terminals of current-sense resistor 2526 are connected to senseinput terminals 2532 and 2533 (one terminal from each of the “sense_I”and “sense_VI” buses) terminals of switching power supply controller1200. The voltage across these sense input terminals is proportional tothe current in CCFL 2525. The pulse-width modulated signals at the gateterminals of MOS switches 2521 and 2522 regulate the power supplied toCCFL 2525.

As can be seen from the above description, each of converters 2570, 2571and 2572 is regulated by a pair of pulse-width modulated drive signals(one from each of “upper_FET_gate” and “lower_FET_gate” groups ofsignals). These signals are preferably non-overlapping (i.e., thesesignals are not both at a high voltage at the same time). For eachconverter, regulation is based on a control loop receiving input signalsrepresenting the values of the controlled variables. FIG. 19 is a blockdiagram summarizing the modules in switching power supply controller1200 providing a control loop for a battery or power supply managementapplication. As shown in FIG. 19, digital to pulse converter module 1201receives from regulation control module (REG) 1204 a 10-bit value,representing the duty cycle of a pulse-width modulated drive signal andaccordingly provides one of seven pairs of pulse-width modulated drivesignals 1901 a and 1901 b. Pulse-width modulated drive signals 1901 aand 1901 b are driven by NFET driver module 1202 (not shown in FIG. 19)as the upper_FET_gate and lower_FET_gate signals from switching powersupply controller 1200. At the same time, seven pair of voltage signals1902 a and 1902 b (i.e., I[0:6], VI[0:6]), each pair representing thevoltages across the terminals of a sense-resistor, are received intosample and hold module 1207. In addition, the voltages at the twoexternal power supply terminals 1903 and 1904 (i.e., Supply A and SupplyB) are also received into sample and hold module 1207. In turn, each ofthese analog voltage signals is sampled and held for conversion byanalog to digital converter 1206, which provides regulation controlmodule (REG) 1204 a 10-bit digital value for each voltage converted.Regulation control module (REG) 1204, which implements a number ofmethods applicable to the control loop, provides digital to pulseconversion module 1201 the appropriate 10-bit value on bus 1907. Ofcourse, the number of bits to be used for each voltage converted valueis a matter of design choice depending upon, for example, the resolutionrequired in the control loop. In fact, within regulation control module(REG) 1204, computation may be carried out at a higher resolution than10-bits and residual values less than the 10-bit resolution may be keptfor multiple cycles to implement specific, higher accuracy controlmethods.

Section 1.2.2.2 Stored External Component Parameters

For each battery or power supply management application, regulationcontrol module (REG) 1204 receives the power supply voltage V_(A) (e.g.,voltage at terminal 2508), the regulated output voltage V_(out) at oneterminal of the current-sense resistor (e.g., voltage at terminal 2531),and the voltage V_(IL) at the other terminal of the current-senseresistor (e.g., terminal 2530). A circuit model of converter 2570 isprovided in FIG. 57.

As shown in FIG. 57, the current I_(L) in the inductor of a buck orboost converter can be determined by the voltage drop across thecurrent-sense resistor 2504 (i.e., V_(out)−V_(IL)) divided by itsresistance R_(sense). Regulation is achieved based upon the measuredvoltage values received and quantities derivable from these receivedvalues. For example, the present invention allows regulation using aninput voltage at terminal 2508 (e.g., the power supply voltage V_(A))and an output current (e.g., current I_(L) in resistor 2504) as controlparameters.

FIG. 26 illustrates the operation of a control loop in accordance withthe present invention. As shown in FIG. 26, such a control loop includesthree phases. At the beginning of a control cycle or cycle, which is a 2microsecond time period in one implementation, the values of controlledvariables (e.g., output voltages) are sampled and digitized in phase2601. Based on these input values, parametric values and an appropriateresponse (e.g., an increase or a decrease in the value of a controlledvariable) are computed at phase 2602. The response (e.g., an increase ora decrease in the duty cycle of the drive signals to the MOS switches)is then applied to effectuate the changes in the controlled variables.

Referring back to FIG. 57, if one takes the parasitic resistances orimpedances of the converter into account, a voltage drop V_(P) can beattributed, for example, to the parasitic resistances R_(on) and R_(L),where R_(on) is the “on” resistance of one of the MOS switches (e.g.,MOS switch 2505 or 2506) and R_(L) is the parasitic series resistance inthe inductor (e.g., inductor 2503). The parasitic resistor R_(on) is notshown in FIG. 57, and the parasitic resistor R_(L) is shown as resistor5701 in FIG. 57. As discussed above, the current I_(L) in the inductorcan be obtained by (V_(out)−V_(IN)) divided by R_(ON)+R_(L).

Voltage V_(P) can be approximated using a method discussed below. Inaddition, a parasitic resistance R_(c) (represented by resistor 5702 inFIG. 57) can be attributed to the parasitic series resistance in theoutput capacitor (e.g., output capacitor 2501). Since the actualinductance L of a commercially available coil may be more than 10%different from its nominal inductance value, and may vary significantlyover the life of the inductor, a method according to the presentinvention allows an accurate inductance value of the inductor to becalculated periodically or on power-up. The present invention alsoprovides a method, described below, for calculating the outputcapacitance C of capacitor 2501.

According to one embodiment of the present invention, regulation of theoutput voltage V_(out) may be carried out in the manner illustrated inFIG. 20. The regulation method illustrated in FIG. 20 recognizes that aninductor has a saturation current I_(LSAT), beyond which the inductorbecomes resistive (i.e., the additional current results in energydissipated as heat and not additional energy being stored in theinductor). Initially, both output voltage V_(out) and inductor currentI_(L) are zero. As shown in FIG. 20, the control method of the presentinvention initially provides a duty cycle in the pulse-width modulateddrive signals to the MOS switches (e.g., close to 100%) that increasesthe current in the inductor at the highest rate until the inductorcurrent reaches a predetermined value of the saturation current I_(LSAT)(e.g., 95% of I_(LSAT)). During this time, the inductor current chargesboth the magnetic field in the inductor and the electric field of theoutput capacitor (e.g., capacitor 2501), so that the inductor currentand output voltage increase along the current/voltage segment 2001. Whenthe inductor current reaches the predetermined current value, asindicated by point 2003, the method of the present invention reduces theduty cycle of the pulse-width modulated drive signals to the MOSswitches sufficiently to maintain the inductor current at that level, sothat substantially all of the energy transferred by the inductor currentis provided to charge the output capacitor. In this portion of thecontrol method, the inductor current and the output voltage followscurrent/voltage segment 2002. When the output voltage reaches thecontrol target voltage V_(target), the control method further reducesthe duty cycle such that the energy transferred by the inductor currentis substantially dissipated in the load (R_(load)) and the parasiticimpedances in the converter. Regulation controls the MOS switch drivesignals such that the inductor current I_(L) and output voltage V_(out)follows current/voltage segment 2004 to an equilibrium value within zone2005.

In some embodiments, when the load is operating in a very low powermode, the duty cycle of the MOS switch drive signals necessary tomaintain output voltage V_(out) within a selected range 2005 centered atV_(target) (“controlled interval”) may fall below a minimum value. Inthat instance, regulation proceeds to an “intermittent” or “cycleskipping” mode in which a pulse may not be sent in each 2 microsecondcycle. Instead, a pulse of fixed duration is sent in the MOS switchdrive signals once every two or more cycles, such that the duty cycleaveraged over the two or more cycles achieves the duty cycle necessaryto maintain the output voltage within the controlled interval.

Subsequently, any power requirement change in a load operation (e.g.,switching on backlighting), would result in a fluctuation in outputvoltage V_(out). The control method adjusts the duty cycle of the MOSswitch drive signals to restore the output voltage V_(out) to theregulated voltage V_(target) along current/voltage segment 2906.

Recalling that the voltage across inductor 2503 is given by:${{\frac{\mathbb{d}V_{out}}{\mathbb{d}t} \cong \frac{\Delta\quad V_{out}}{\Delta\quad t}} = \frac{I}{C}},$and because along current/voltage segment 2001 the change in current ΔIin inductor 2503 can be approximated by the change in voltage dropΔ(V_(out)−V_(IL)) across sense-resistor 2504 divided by its resistanceR_(sense) over a 2 microsecond cycle (Δt), a first order approximationof the inductance L of inductor 2503 can be thus calculated. Similarly,since the change in output voltage ΔV_(out) over a 2 microsecond frame(Δt) is given by${{L\frac{\mathbb{d}i}{\mathbb{d}t}} \cong {L\frac{\Delta\quad I}{\Delta\quad t}} \cong {V_{IL} - V_{in}}},$and because, along current/voltage segment 2002, the voltage drop(V_(IL)−V_(in)) across inductor 2503 is approximately zero$\left( {{i.e},\quad{{L\frac{\mathbb{d}i}{\mathbb{d}t}} \cong {L\frac{\Delta\quad I}{\Delta\quad t}} \cong 0}} \right)$and the current I is given by the voltage (V_(out)−V_(IL)) acrosssense-resistor 2504 divided by its resistance R_(sense), capacitance Ccan also be calculated under no-load conditions, or the combinedcapacitance of C and C_(LOAD). During the same time interval, theparasitic resistance R_(L) of inductor 2503 can be approximated from thecurrent I_(L) and the small voltage drop (V_(IL)−V_(in)) across inductor2503 by (V_(IL)−V_(in))/I_(L).

Other control parameters can be similarly calculated. For example, in abuck converter, the efficiency E at any given time is given by${E = \frac{V_{out}}{V_{i\quad n}D}},$where D is the duty cycle of the MOS switch drive at the current time.Efficiency E can be updated at the beginning of each cycle.

At operating point 2005, before the load is applied, the ripple voltageat the output and the parasitic series resistance R_(C) of the outputcapacitor can be approximated using the ratio of the ripple outputvoltage divided by the average inductor current, which is determined bycalculating a weighted average of the maximum and minimum inductorcurrents over a 2 microsecond cycle. The maximum and minimum inductorcurrents are obtained by the difference in the maximum and minimumvoltage drop (V_(out)−V_(IL)) across current-sense resistor 2504,sampled respectively immediately prior to MOS switch 2505 opening andclosing, respectively. The weighted average of the maximum and minimumcurrents is calculated by weighting the maximum and minimum currents bythe duty cycle D. In other words, as illustrated in FIG. 21:${\overset{\_}{I}}_{L} = \frac{{D\left( {V_{out} - V_{IL}} \right)}_{\max} + {\left( {1 - D} \right)\left( {V_{out} - V_{IL}} \right)_{\min}}}{2}$

The parasitic series resistance R_(C) of the output capacitor can beapproximated by the difference ΔV_(out) of the output voltage ripple,divided by the average current Ī_(L), or${R_{C} = \frac{\Delta\quad V_{out}}{{\overset{\_}{I}}_{L}}},$whereΔV _(out)=(V _(out max) −V _(out min)).

In the controlled interval 2005, while the inductor current isapproximately constant, the voltage drop between the input voltageV_(in) and the voltage V_(IL) 2530 at the common terminal of theinductor and the current-sense resistor 2504 represents the voltage dropacross the parasitic resistance of one of MOS switches 2505 and 2506 andthe parasitic series resistance of inductor 2503.

The efficiency loss due to switching at an MOS switch can beapproximated using data obtained from a temporary change in thepulse-modulated MOS switch drive signals. This temporary change isillustrated in FIG. 22. FIG. 22 shows a pulse-modulated MOS switch drivesignal over two intervals A and B of equal durations and equal dutycycles. (Although FIG. 22 shows interval A to be only two cycles wide,in practice, the number of cycles in each interval should be a highernumber to increase accuracy.) However, the on-portion of each cycle ininterval A is shorter than the on-portion of each cycle in interval B,so that the number of on-pulses in interval A is much higher than thecorresponding number of on-pulses in interval B for the same duty cycle.Thus, any difference in efficiencies in the converter between theseintervals can be attributed to switching losses in the MOS switches dueto each switch's parasitic impedances. Any difference ΔV_(out) in theoutput voltage is dissipated across the parasitic resistance R_(p).,which is the sum of resistance R_(on) and the resistance R_(L) ofinductor 2501. R_(p) can therefore be estimated by${R_{p} = \frac{\Delta\quad V_{out}}{\overset{\_}{I}}},$where Ī is the average current over intervals A and B.

FIG. 57 also shows input filter capacitor 5705 which has a parasiticresistance R_(cin) (represented by resistor 5704), and an input powersupply (represented by battery 5706) having a parasitic resistance R_(s)(represented by resistor 5703).

Section 1.2.2.3 Absolute Value Power Supply Control Loop, IncludingUsing Control Dead Band and Variable Gain

The calculated parameter values, including the inductance, the outputcapacitance and the parasitic resistances, can be used to implement thecontrol methods. In the prior art, power regulation is usually providedby an adaptive feedback mechanism in which the corrective change inpulse-width modulation is a linear function of the error. The error isthe difference between the actual and target values of the controlledvariable which, in this instance, is the regulated output voltageV_(out). Under one method, called the “PID” method, the feedbackcorrection is a “proportional feedback” represented by a linear sum ofconstant multiples of the error, a derivative of the error, and anintegral of the error. In such a PID adaptive system, the systemfunction H'(s) of an adaptive system is given by:${{H^{\prime}(s)} = \frac{H(s)}{1 - {\left( {k_{1} + {k_{2}s} + \frac{k_{3}}{s}} \right){H(s)}}}},$where k₁, k₂, and k₃ are constants and H(s) is the open-loop systemfunction. In most systems, because the error is small, the feedbackcorrection is dominated by the integral of the error, and it may take alarge number of cycles to restore the regulated voltage to V_(target).According to the present invention, however, a predictive technique isused. Under a predictive technique, the amount of correction is obtainedby calculating an estimated restorative change in the control parameters(e.g., duty cycle input) that is needed to correct the error.

A control method according to one embodiment of the present invention isillustrated by flow diagram 2300 of FIG. 23. As shown in FIG. 23, atstep 2301, the control method examines an output voltage error value,which is given by error=V_(out)−V_(target). If the error value is lessthan a predetermined threshold value (“dead band”), no adjustment isdeemed necessary, and the method returns to step 2300. Otherwise, i.e.,if the value of error exceeds the threshold, a current-constrained dutycycle D_(i) is computed at step 2302. Current-constrained duty cycleD_(i) represents the duty cycle that would bring the inductor currentI_(L) to within a predetermined offset (e.g., 0 amps) of its saturationvalue I_(LSAT). As explained above, the control method should not driveMOS switches 2505 and 2506 beyond this duty cycle.

Current-constrained duty cycle D_(i) satisfies the equation:${{{L\frac{\mathbb{d}I_{L}}{\mathbb{d}t}} \cong {L\quad\frac{\left( {I_{LSAT} - I_{L}} \right)}{D_{i}T}} \cong V_{L}} = \left( {V_{out} - V_{i\quad n} - V_{p}} \right)},$where V_(L) is the voltage across inductor 2503, T is the cycleduration, and V_(p) is the total voltage drop across the parasiticresistances of inductor 2503 and MOS switch 2505 and the resistance ofcurrent-sense resistor 2504. The expression$\frac{\left( {I_{LSAT} - I_{L}} \right)}{D_{i}T}$represents the approximate rate of change of inductor current necessaryto bring the inductor current I_(L) to saturation current I_(LSAT) overthe time period D_(i)T (i.e., when the drive signal of MOS switch 2505is “on”. Solving this equation, we obtain the current-constrained dutycycle D_(i):${D_{i} = {{\frac{L}{T\left( {V_{out} - V_{i\quad n} - V_{p}} \right)}\left( {I_{LSAT} - I_{L}} \right)} \cong {K\left( {I_{LSAT} - I_{L}} \right)}}},$where we have defined K to be the value$\frac{L}{T\left( {V_{out} - V_{i\quad n} - V_{p}} \right)}.$Observing that the product of K and inductor current I_(L) provides theduty cycle, the value of K is temporary stored and reused withoutrecomputation for calculating the voltage-based duty cycle D_(V), whichis calculated next in step 2303.

Voltage-based duty cycle D_(V) is the duty cycle that is necessary tocorrect for the output voltage V_(out) by the value of error (i.e.,bringing the output voltage V_(out) to the target voltage V_(target)),without regard for the increase in inductor current I_(L). For a givenefficiency E, the nominal duty cycle D_(nom) necessary for providing anoutput voltage of V_(target) is given by$D_{nom} = {\frac{V_{target}}{V_{i\quad n}*E}.}$The voltage-based duty cycle D_(V) is the sum of nominal duty cycleD_(V) and an adjustment ΔD_(v) to this nominal duty cycle.

Using the equation derived above for the parasitic series resistanceR_(C) of the output capacitor, the additional current ΔI_(L) needed tocorrect for the value of error is given by:${{\Delta\quad I_{L}} = {- \frac{error}{R_{C}}}},$and the incremental change ΔD_(v) in duty cycle D_(v) is given${\Delta\quad D_{V}} = {{K\quad\Delta\quad I_{L}} = {- {\frac{K*{error}}{R_{C}}.}}}$Thus, voltage-based duty cycle D_(V) is given by$D_{V} = {{D_{nom} + {\Delta\quad D_{V}}} = {D_{nom} - {\frac{K*{error}}{R_{C}}.}}}$

At step 2304, the lesser one of the current-constrained duty cycle D_(i)and the voltage-based duty cycle D_(V) is selected, so that theresulting duty cycle does not provide a current exceeding the inductorsaturation current I_(LSAT). In some embodiments, this selected dutycycle also cannot fall below a predetermined minimum. The selected dutycycle is then applied to the MOS switch drive signals. The drive signalsfor the primary MOS switch (i.e., MOS switch 2505 connecting the powersupply voltage to the inductor) and the secondary MOS switch (e.g., MOSswitch 2506) are made non-overlapping. Control method 2300 returns tostep 2301.

According to another embodiment of the present invention, a controlalgorithm seeking to restore the converter back to equilibrium operationcan be implemented. Equilibrium is achieved when the followingconditions are met: (a) the output voltage V_(out) of circuit targetvoltage equals the target voltage V_(target), (b) the average currentĪ_(L) in inductor 2503 equals the average current Ī_(load) drawn by theload, and (c) no change in inductor current between the DPC frames.

The algorithm uses a linear approximation of a change in inductorcurrent over a time period that is much less than the periods of thefundamental frequencies of the converter. Within a DPC frame of durationT, the primary switch (e.g., switch 2505) is “on” (closed) for aduration T_(P), and the secondary switch (e.g., switch 2506) is “on”(closed) for a duration T_(S). Thus, an incremental current ΔI_(L)results in a change in output voltage V_(out). The incremental currentΔI_(L) can be approximated by time-weighting components incrementalinductor currents ΔI_(L(P)) and ΔI_(L(S)), corresponding to the timeperiods in which the primary and the secondary switches are “on”,respectively. ΔI_(L(P)) and ΔI_(L(S)) are given by: $\begin{matrix}{{\Delta\quad I_{L{(P)}}} = {{T_{P}\left( \frac{V_{i\quad n} - {Vp} - V_{out}}{L} \right)} = {T_{P}\left( \frac{V_{i\quad n} - {I_{L}R_{pp}} - V_{out}}{L} \right)}}} \\{{\Delta\quad I_{L{(S)}}} = {{T_{S}\left( \frac{0 - {Vs} - V_{out}}{L} \right)} = {T_{S}\left( \frac{{{- I_{L}}R_{ss}} - V_{out}}{L} \right)}}}\end{matrix}$

where V_(P) and V_(S) are the voltage drops across the resistancesR_(pp) and R_(ss) in the current paths during durations T_(P) and T_(S),respectively. Referring to FIG. 57, R_(pp) and R_(ss) are given by:$R_{pp} = {R_{sense} + R_{L} + R_{p} + \frac{R_{source}R_{cin}}{R_{source} + R_{cin}}}$R_(ss) = R_(sense) + R_(L) + R_(s)  

In one implementation, the sense resistor may be omitted (i.e.,R_(sense)=0). In that implementation, inductor current I_(L) is notmeasured but estimated. Because${V_{out} = {V_{C} + {\left( {I_{C} + {\Delta\quad I_{C}}} \right)R_{C}} + \frac{\Delta\quad V_{C}}{2}}},$and ${{\Delta\quad V_{C}} = \frac{I_{C}T}{C}},$ΔI_(L) is given by: $\begin{matrix}{{\Delta\quad I_{L}} = \frac{{\frac{T_{P}}{T}\left( {V_{in} - {R_{pp}I_{L}}} \right)} - {\frac{T_{S}}{T}R_{SS}I_{L}} - \left( {V_{C} + {I_{C}\left( {\frac{T}{2C} + R_{C}} \right)}} \right)}{\frac{L}{T} + R_{C}}} & (1)\end{matrix}$

where I_(C)=I_(L)−I_(load)

Substituting${{D\quad C} = {\frac{T_{P}}{T_{P} + T_{S}} = \frac{T_{P}}{T}}}\quad$and$\left( {1 - {D\quad C}} \right) = {\frac{T_{S}}{T_{P} + T_{S}} = \frac{T_{S}}{T}}$into equation (1) and solving for the duty cycle DC to obtain:$\begin{matrix}{{D\quad C} = {\frac{{\Delta\quad{I_{L}\left( {\frac{L}{T} + R_{C}} \right)}} + {R_{SS}I_{L}} + V_{C} + {I_{C}\left( {\frac{T}{2C} + R_{C}} \right)}}{V_{i\quad n} + {I_{L}\left( {R_{ss} - R_{pp}} \right)}} = \frac{{\Delta\quad{I_{L}\left( {\frac{L}{T} + R_{C}} \right)}} + {R_{SS}I_{L}} + V_{out}}{V_{i\quad n} + {I_{L}\left( {R_{ss} - R_{pp}} \right)}}}} & (2)\end{matrix}$

Equation (2) can be used to predict a duty cycle DC that should beapplied to provide a given incremental inductor current ΔI_(L). Theincremental current to be used is selected from a number of applicableincremental inductor currents (i.e., ΔI_(L)'s)

FIG. 58 is a flow chart illustrating the steps of control algorithm5800, in accordance to with one embodiment of the present invention. Asshown in FIG. 58, at step 5801, input voltage V_(in) and output voltageV_(out) of the power converter are sampled, and inductor current I_(L)is estimated using equation (1):I _(L)(t _(n))=I _(L) (t _(n−1))+ΔI(t _(n−1))

At step 5802, the load current I_(load) in the current DPC frame is thencalculated. The expected incremental current ΔI_(L)(t_(n)) in thepresent DPC frame charges capacitor 2501, raising its voltage V_(C) byan amount ΔV_(C). The load current I_(Load) can be estimated by:${I_{load}(t)}_{n} = {{{I_{L}\left( t_{n} \right)} + {\Delta\quad{I_{L}\left( t_{n} \right)}} - {I_{C}\left( t_{n} \right)}} = {{I_{L}\left( t_{n} \right)} + {\Delta\quad{I_{L}\left( t_{n} \right)}} - \frac{C\left( {{\Delta\quad{V_{out}\left( t_{n} \right)}} - {R_{C}\Delta\quad{I_{L}\left( t_{n} \right)}}} \right)}{T}}}$

where ΔV_(out), (t_(n))=V_(out)(t_(n))−V_(out)(t_(n−1))

(From this point forward, the parameter values are understood to bethose of the current DPC frame, and the designation t_(n) is dropped forclarity.)

At step 5803, the voltage on output capacitor 2501 can also beestimated, at the end of the current frame, to be:$V_{C} = {V_{out} + {I_{C}\left( {\frac{T}{C} - R_{C}} \right)}}$

The change in capacitor voltage over the following DPC frame can also beestimated: ${\Delta\quad V_{C}} = \frac{\Delta\quad I_{C}T}{C}$

At step 5804, a number of different incremental inductor current valuescan be calculated as candidates which corresponding duty cycles can beused to restore the converter towards equilibrium operation. To restoreoutput voltage V_(out) back to target voltage V_(target) in the next DPCframe, assuming no further change in load current I_(load), an inductorcurrent change ΔI_(L(target)) is required:${\Delta\quad I_{L{({target})}}} = {I_{load} - I_{L} + \frac{C\left( {V_{target} - V_{C} - {\Delta\quad V_{C}}} \right)}{T}}$

However, ΔI_(L(target)) may be so large in magnitude that it requires aphysically impossible duty cycle—i.e., a duty cycle that is greater than100% or less than 0%. However, if a 100% duty cycle (i.e., T=T_(P) andT_(S)=0) is to be applied in the next DPC frame, the incremental currentΔI_(L(100%)) is:${\Delta\quad I_{L{({100\%})}}} = \frac{{\frac{T_{P}}{T}\left( {V_{i\quad n} - {R_{pp}I_{L}}} \right)} - \left( {V_{C} + {I_{C}\left( {\frac{T}{2C} + R_{C}} \right)}} \right)}{\frac{L}{T} + R_{C}}$

Similarly, if a 0% duty cycle (i.e., T=T_(S) and T_(P)=0) is to beapplied in the next DPC frame, the incremental current ΔI_(L(100%)) is:${\Delta\quad I_{L{({0\%})}}} = \frac{{{- \frac{T_{S}}{T}}R_{SS}I_{L}} - \left( {V_{C} + {I_{C}\left( {\frac{T}{2C} + R_{C}} \right)}} \right)}{\frac{L}{T} + R_{C}}$

Incremental current ΔI_(L(0%)) is amount of inductor current that can bewithdrawn from the converter in a DPC frame. In this instance, controlalgorithm 5800 computes a quantity “maximum current” ΔI_(L-MAX) that canbe removed in one DPC frame to achieve I_(load)=I_(L):ΔI _(L-MAX) =I _(load) −I _(L) −ΔI _(L(0%)))

Further, at any given time, the increase in inductor current is limitedby the saturation current I_(L-SAT) of the inductor:ΔI _(L-SAT) =I _(L-SAT) −I _(L)

Thus, at step 5805, the predictive control algorithm chooses as ΔI_(L)the minimum of ΔI_(L(target)), ΔI_(L(100%)), ΔI_(L-MAX) and ΔI_(L-SAT).Plugging the chosen ΔI_(L) into equations 2 yields at step 5806 a dutycycle between 0% and 100%, inclusive, to correct the transient conditionat the output. The duty cycle corresponding to the selected ΔI_(L) isthe used at step 5807 to control the primary and secondary switches inthe next DPC frame. The selection of ΔI_(L) discussed above is valid ifthe system is recovering from a transient caused by an increase inI_(load). If I_(load) is decreased a similar flow results with slightchanges in the equations. In this caseΔI_(L-MAX)=I_(load)−I_(L)−ΔI_(L(100%))) and ΔI_(L-SAT)=−I_(L).

The predictive control algorithm will now choose the maximum ofΔI_(L(target)), ΔI_(L(0%)), ΔI_(L-MAX) and ΔI_(L-SAT).

FIG. 59 illustrates the operation of control algorithm 5800 of FIG. 58.In FIG. 59, the output voltage V_(out) of a power converter is shown aswaveform 5901, the inductor current I_(L) is shown as waveform 5902, andthe duty cycle of the primary switch is shown as waveform 5903. Prior totime t=0, the converter operates at 50% duty cycle, the input voltageV_(in) is at 8 volts, and the output voltage V_(out) is at 4 volts, andthe load current I_(load) and the inductor current are both zero. Attime t=0, a load is connected to the output terminals of the powerconverter, causing an increase of load current by 2 amperes. At thistime, a dip in the output voltage is detected, leading to a non-zeroestimate of a change in inductor current of 0.059 amperes over theinductor current of the last DPC frame (i.e., the converter deviatesfrom equilibrium). This change in inductor current results in anestimated non-zero load current I_(load) of 1.022 amperes, and an outputcapacitor voltage V_(C) of 3.850 volts. In this instance, I_(LSAT) is 3amperes. To restore the converter back to equilibrium, control algorithm5800 estimates ΔI_(L(target)), ΔI_(L(100%)), ΔI_(L-max) and ΔI_(LSAT) tobe 5.882, 0.879, 2.630 and 2.941 amperes, respectively. Thus, at timet=2 microseconds (us), 100% duty cycle is applied (corresponding toΔI_(L(100%))=0.823). As the estimate of the load current is low, and thecorrection applied is small, the sampled output voltage V_(out)continues to drop.

At time t=2 us, the sampled output voltage dropped to 3.703 volts, butthe estimated inductor current I_(L) rises to 0.059+0.879=0.938 amperes,the load current is estimated to be 2.378 amperes, and the capacitorvoltage is predicted to drop to 3.606 volts. As in the previous cycle,control algorithm 5800 selects a 100% duty cycle for a ΔI_(L(100%)) of0.870 amperes. In the next 2 cycles (t=4, 6 us), control algorithmmaintains a 100% duty cycle operation to ramp the estimated inductorcurrent I_(L) to 2.651 amperes. At this inductor current, the sampledoutput voltage V_(out) drops to 3.505 volts, but the inductor currentI_(L) is now sufficient to prevent further dropping.

At time t=6 us, control algorithm 5800 estimates ΔI_(L(target)),ΔI_(L(100%)), ΔI_(L-max) and ΔI_(LSAT) to be 3.893, 0.802, 0.102, and0.349 amperes, respectively. Thus, a duty cycle of 55.03% is selectedfor the next DPC frame (i.e., time t=6 us to t=8 us). At times t=8, 10,12, and 14 us the control algorithm selects successive values ofΔI_(L-max), i.e., 0.020, 0.016, 0.017, and 0.016 amperes, respectively,corresponding to duty cycles of 51.03%, 51.79%, 52.93%, and 53.97%. Att=14 time, the sampled output voltage V_(out) climbs back to 3.728volts.

At time t=16 us, control algorithm 5800 calculates that the targetvoltage can be achieved with an incremental inductor current of −0.205amperes, corresponding to a duty cycle of 40.76%. At time t=18 us, thesampled output voltage V_(out) achieves 3.89 volts, and predicted toachieve 4.0 volts at the end of the DPC frame. A 14.70% duty cycle isselected to maintain the output voltage V_(out) and remove inductorcurrent to achieve the equilibrium condition that inductor current Ī_(L)equals load current Ī_(load). At time t=22 us, the sampled voltageV_(out) achieves 4.00 volts and inductor current Ī_(L) equals loadcurrent Ī_(load), control algorithm 5800 selects a 52.68% duty cycle,which approaches the long term equilibrium duty cycle of 53.01%. Thelong term equilibrium duty cycle of 53.01% is achieved at time t=24 us.In one embodiment, a “dead band” is provided in the vicinity of thetarget output voltage V_(target) within which the voltage output V_(out)can drift uncorrected. Correction occurs when the output voltage V_(out)drifts outside of the dead band. The dead band renders the regulationless susceptible to high frequency noise.

In another embodiment, a high voltage limit and a low voltage limit areprovided on either side of the target voltage and the dead band. Shouldoutput voltage V_(out) drift outside of the dead band, but within theband defined by the high limit and the low limit, a fixed correctionamount is applied without taking regulation control module (REG) 1204out of low power mode to restore output voltage V_(out) to within thedead band. Algorithms requiring computation; such as control algorithm5800 described above, are used only when the correction exceeds the banddefined by the higher and lower limits. In this manner, power regulationcan occur for a substantial amount of time under low power mode.

In another embodiment, the loop gain is made less than one (i.e., only afraction of the value of error is corrected) and varies based upon theabsolute value of error. For example, in one embodiment, for largeoutput voltage excursions (e.g., error is greater than 0.5 volts), ahigher loop gain is provided (e.g., 80%). For a smaller output deviation(e.g., less than 0.05 volts), a smaller (e.g., 20%) or zero loop gaincould be used. Variable loop gain allows a quick response to largechanges in the power requirements at the load, but provides higherimmunity to high frequency noise for small deviations.

Section 1.2.2.4 Maximum/Minimum Bounded Control Loop

In another embodiment, a minimum duty cycle, a maximum duty cycle, orboth, can be provided. The minimum or maximum duty cycle limits theconverter between a minimum power or a maximum power. In a system inwhich several converters draw power from the same power source, such asconverters 2570 and 2571 shown in FIG. 25, limiting each converter to amaximum power prevents a large power excursion in one converter frominterfering with the operation of another converter. For example, if amomentary large current is drawn from converter 2570 (FIG. 25), withouta maximum duty cycle limit, a voltage dip may occur at power supply A(terminal 2508), which is also connected to converter 2571. A largevoltage dip in power supply A may cause a transient response atconverter 2571. A maximum duty cycle limit may prevent suchinterference. The prescribed minimum and maximum duty cycle limits canbe stored in registers and made programmable by the users.

Section 1.2.2.5 Low Frequency Closed/High Frequency Predictive ControlLoop

As discussed above, for each cycle, sampling and digitizing inputvalues; response computation; and response application must all beaccomplished within the cycle. In one embodiment of the presentinvention, to allow an optimal response to be implemented withoutrequiring an extension to the duration of the cycle to accommodate theadditional computation required, a control method uses open-loopregulation for certain power events that are known and characterized àpriori. For example, in a dynamic random access memory (DRAM) system, arefresh event—during which the storage cells of DRAM system aresystematically read—occurs in approximately regular intervals. For sucha known power event, the power requirement (“signature”) of the eventand an appropriate response can be characterized and stored in memory.When the known power event occurs and is recognized, the pre-computedresponse can be applied in the current and subsequent cycle withoutrecomputation (hence, “open-loop”). FIG. 24 and FIG. 27 illustrate alow-frequency closed loop and high-frequency open loop control method,according to one embodiment of the present invention.

As shown in FIG. 24, at step 2401, a state machine is provided to searchfor the signature of a recognized power event from the sampled inputvalues of controlled variables. This state machine may search for apower event signature, for example, from a table of values of controlledvariables stored over a moving window of several cycles. FIG. 27 showswaveform 2701, which represents a signature of a power event exhibitedin the input values of a controlled variable (e.g., output voltage),occurring between time t₀ and time t₆. Before a signature is recognized,the state machine selects at step 2402 a closed loop control method forpower regulation, such as any of the controlled loops discussed above.For example, referring to FIG. 27, between cycle t₀ and cycle t₁, whenthe signature of the power event is recognized, the closed loop methodis used to provide the response (in this case, an incremental change induty cycle of the MOS switch drive signals). Thus, the portion ofwaveform 2702 between cycle t₀ and cycle t₁—a period of perhaps severalcycles—represents the response resulting from the operation of a closedloop control method. However, at cycle t₁, the state machine recognizesthe power event, and switches the control method to the open loopcontrol method at step 2404 for the time period between cycles t₁ andt₆. The open loop method is programmed to deliver the incremental dutycycles shown in waveform 2702 between frames t₁ and t₆, so long as thesampled values of the controlled variables during the same period matchthe stored expected values. Otherwise, i.e., if the input values of thecontrolled variables during operation of the open loop control methodare not what are expected, the state machine reverts to the closed loopmethod at step 2402. In addition, as shown in FIG. 24, the closed loopmethod can also operate in concert with the open loop method to correctany residual error value that is not corrected by the open loopresponse.

A simplification of this embodiment may occur when a predictable, knownpower event (e.g., sounding an alarm buzzer or an excitation event by aCCFL supply) occurs wherein the event timing is known (since it is underprogram control) and the impact is known, hence appropriate power supplyresponse is known ahead of time. Here one does not have to recognize thesignature but simply impose the appropriate response. It can be viewedas this same control means but with the special case wherein the time torecognize the “signature” is zero.

In one embodiment, the closed loop response integrates an error valueover multiple cycles and provides a corrective response only when theaccumulated error exceeds a threshold or when a predetermined number ofelapsed number of cycles is exceeded. Alternatively, the incrementalduty cycle corrections can also be accumulated over multiple cycles andapplied only when the accumulated corrections exceed a threshold or whena predetermined number of elapsed number of cycles is exceeded. In thismanner, the closed loop method acquires immunity to high frequency noiseand provides a dampened transient response.

Section 1.2.2.6A Sequential Transient Recovery Algorithm

In one embodiment, multiple signatures can be searched simultaneously.To handle the case of a sequential occurrence of multiple transients,i.e., when a recognized power event occurs before another recognizedpower event completes, the state machine searches linear super-positionsof two or more power event signatures offset in time to detect such anoccurrence. FIG. 28 illustrates a sequential transient recovery controlmethod, in accordance with one embodiment of the present invention. InFIG. 28, waveforms 2801 and 2802 represent the signatures of two powerevents occurring singly at times t₀ and t₁, respectively. After theoccurrence of the power event of waveform 2801 is detected at time to,the state machine also searches composite signatures, such as waveform2803, which is the linear super-position of waveforms 2801 and 2802offset by time interval (t₁−t₀). Should the power events of waveform2801 and 2802 occur, the values of the controlled variables match one ofthe composite signatures. An appropriate response corresponding to alinear super-position of the corresponding programmed responses can beapplied as an open loop correction.

Section 1.2.2.6 Adjacent Phase Sampling

In switching power converters, it is necessary to measure voltage andcurrent being provided to make adjustments, if needed. In typical priorart solutions, the entire waveform is applied to an error amplifier andthen subsequently to a comparator. The error amplifier, if nothing weredone, would faithfully amplify the output voltage ripple. This wouldpresent a rippled signal to the comparator and result in unstable orunpredictable information being used by the comparator to set the pulsewidth modulation. This is unacceptable, so the typical prior artapplication using an error amplifier uses filtering sometimes calledcompensation in the error amplifier's feedback loop, to filter out thevoltage ripple, resulting in an average value being applied to thecomparator. There are at least two undesirable consequences to this.First of all, it is the average value of the ripple voltage that isregulated. Secondly, the frequency response of this error amplifier hasbeen dramatically reduced over what it might have been, reducing theperformance of the power supply.

An advantage is provided in the present invention by using synchronoussampling. By sampling the voltage at the same point in time relative tothe switching waveform, the switching ripple itself is removed. Morethan that, it is possible to choose a point that corresponds to acritical parameter. A very common critical parameter in power converterdesign is to maintain a minimum voltage such that this voltage is abovethe minimum requirements of the load. Microprocessors and memories arevery intolerant of voltage excursions below this minimum point. Bysampling at the voltage minimum, then, a very stable non-ripplingvoltage measurement is developed over time for the voltage minimumproduced by the regulator. This permits the regulator to deliver theminimum voltage required by the microprocessor. In a prior art solution,it is necessary to margin the regulation point to account for theexcursions above and below the average, which was derived when the erroramplifier's frequency response was compensated and the voltage rippleaveraged (filtered). Similarly, on current, by taking measurements ofthe current not just synchronously with a switching waveform to removethe current ripple effect in the measurement, but also at a point wherethe current might be at a maximum and other advantage is achieved. Themaximum current is of interest because it is needed to prevent the coilfrom saturating; that is, when the current goes above the maximum ratingfor a coil, it no longer is able to act as an inductor, with its coilinductance reduced, the very small resistance of the wire in the coilbecomes the dominate parameter, and the current can increase rapidly, aswell as creating serious noise problems in terms of radiated H and Efield noise from the coil and associated circuitry. Minimizing themaximum of peak current also minimizes the amount of noise that can comein from an external supply. Controlling the maximum peak current can beused to regulate the rate at which a power converter turns on the rampof its voltage. So the benefits of synchronously sampling voltage andcurrent are twofold. First, the elimination of the switching noise, theripple in the voltage and the current in the measurement are removed bydefinition and, secondly, it allows the circuit to precisely regulatethe critical portion of the parameter, rather than an average of theparameter which would result in the circuit attempting to infer wherethat critical point occurs.

The switching power converter system according to the present inventionuses sample data techniques to remove the switching noise of eachindividual switching power supply by, for example sampling the outputvoltage at an adjustable point or a fixed point immediately before therising edge of the gate drive voltage to the upper transistor in a buckconverter configuration. Consider buck converter 1301.2 in FIG. 46, andrefer to FIG. 42. The signal shown in the latter figure is the gatedrive signal to FET QT₁. The reference characters I and V_(o) in FIG. 42are provided to indicate the time at which output voltage V_(o) and thecurrent I are measured with respect to the gate drive voltage applied totransistor QT₁. That is, V_(o) is measured before transistor QT₁ beginsto conduct current into inductor L1. By measuring V_(o) at this time,output voltage V_(o) is at the lowest point in the switching waveform,and therefore it has stabilized from the prior switching cycle. Bymeasuring the current through the inductor L1 just prior to FET QT₁ceasing conduction, the peak current in the inductor L1 is measured oneach switching cycle. Similarly, the current is measured at a point atwhich it has been allowed the most time to ramp towards its final value,i.e., just before FET QT₁ turns off.

As illustrated in FIG. 42, in the successive switching cycles, the firstof which extends from A to C, and the second of which extends from C toE, V₀ and I are measured at the same relative position with respect tothe gate drive voltage. Points A and C in the waveform indicate theleading edge of the gate drive voltage for transistor QT₁. Although inFIG. 42 the voltage measurement time and the current measurement timeare illustrated as being immediately before the leading edge andtrailing edge, respectively of the gate drive voltage to transistor QT₁,other locations may be used. By measuring at the same relativepositions, ripple in the voltage and current are removed.

FIG. 42A illustrates a typical boost switching power converter. In thiscircuit, FET B.2 is turned on for a first time period and FET U.2 isturned off during the first time period. The current I through inductorL1.2 flows in the direction indicated. During a second time period FETB.2 is turned off and FET U.2 is turned on, resulting in a voltage beingstored in capacitor C1.1. In accordance with the present invention, in aboost circuit configuration such as that shown in FIG. 42A, controlpulses are periodically applied to the gate of FET B.2, and the outputvoltage V_(o) is measured immediately before FET B.2 begins to conduct.The current I may be measured immediately before the gate drive signalto FET B.2 is terminated. Following the foregoing procedure with a boostconverter provides the same advantages as those pointed out above for abuck converter. In one embodiment of the present invention, theswitching cycle is 2 usec, however other cycle times may be used. Anexemplary sample time for measurement is 2 nseconds. Thus the sampletime is a very small portion of the entire cycle, which is in contrastto the prior art in which the parameter of interest is measuredcontinuously.

The operation of sample and hold circuit 1207 is described in detail inother portions hereof. Briefly, however, commands from digital pulsecontrol wrapper 1201 instruct the sample and hold circuit 1207 when totake samples of the voltage (V₀) and the current (I). Software inregulation control module (REG) 1204 determines, based on themeasurements, what is needed to return the supply which is beingregulated to equilibrium. This determination may be made by acomputation, or by a look-up table. The process of returning the supplybeing regulated to equilibrium means bringing V₀ to the target voltagefor the supply; making the average current in the inductor equal to theload current; and making the change in current over the switching cyclezero. Based on the determination of what is required to bring theregulated supply to equilibrium, output signals from regulation controlmodule (REG) 1204 to digital pulse control wrapper 1201 instruct thedigital pulse control wrapper 1201 the width of the gate drive pulseswhich should be sent to both transistors.

If a single switching power converter is being controlled, the foregoingwould be sufficient to remove the noise of that single switching powersupply. That is, its own switching noise would not influence its voltageand current measurements being used to regulate its output voltage. Inone embodiment of the present invention, there are seven power supplies.If the gate drive signals to the switching transistors were appliedwithout reference to each other, it is probable that at some pointchannel-to-channel interference could occur. For example, if the senselines of a first channel, which we'll call channel Ø, were close to thesense lines of a second channel, which we'll call channel 1, channel 1might switch where channel Ø is attempting to take its voltagemeasurement and corrupt that voltage measurement. If no coordinationbetween switching times were to be provided, because of the randomnature of where the channel 1 switching might occur, this interferencemight be for a single cycle, or several, or random in fashion. In thepresent invention, all of the switching signals are derived internally,i.e., they're all running off of the same clock and they are allscheduled by logic. Channel-to-channel interference is avoided bycarefully scheduling the switching points of each individual phase. Inthe above example, were channel 1 to have its gate drive signal appliedat the time when the sample was being taken from channel Ø interferencecould affect the measurement. In accordance with the present invention,the switching times are rescheduled as shown in FIG. 42B. As will beappreciated by reference to FIG. 42B, by rescheduling the gate drivesignal to the transistor in CH1 to a later time, the measurements ofvoltage and current in CHØ will not be affected by the switching in CH1.Similarly, the phase of each switching waveform for each power supplycan be changed so that no phase interferes with any other channel. Inthis fashion, the switching noise is removed from the samples bysampling synchronously within each channel with its own switching andthen across channels by scheduling the phase of each adjacent channel.The scheduling of the gate drive waveform is performed by the edgescheduler portion of regulation control module (REG) 1204 which isdescribed above.

Section 1.2.2.7 Stored External Component Parameters

Central processing module (SYS) 1205 may be programmed by a host withthe operating parameters of the switching power converters to becontrolled. For example, the desired voltage levels, expected powersdrawn by loads, and circuit values such as the inductance andcapacitance values of components and the operating characteristics ofthe transistors associated with the switching power supplies may all bestored in non-volatile memory. As described with respect to regulationcontrol module (REG) 1204, central processing module (SYS) 1205 may usethese external parameters to calculate an expected pulse width for eachswitching power supply prior to normal operation. Using these expectedpulse widths, central processing module (SYS) 1205 schedules thecorresponding pulse edges across a DPC cycle and provides the pulse edgeschedule to regulation control module (REG) 1204. By beginning normaloperation with this pulse edge schedule, the rescheduling of pulse edgesrequired of regulation control module (REG) 1204 during normal operationmay be reduced.

Section 1.2.2.11 Dynamic Synchronous and Non-synchronous Operation

Referring to FIG. 29, buck converter 49 is illustrated and should bereferred to in connection with the following operational explanation. Asillustrated in FIG. 29, a 10V input is provided and the transistors areswitched appropriately to provide a 5V output V₀. If it is desired toprovide an output voltage which is closer to the input voltage, then acurrent curve such as that illustrated in waveform A in FIG. 43 would beseen through inductor L50. More particularly, a current I_(max) isreached during a period when FET 50 is conducting, and the current uponthe cessation of conduction FET 50 falls during the conduction of FET51, moving during the operational cycle to I_(min). This cycle isindicated in waveform A by the reference character CYCLE.

Curve A shows that current is always flowing in the inductor L50, nevergoing to 0 hence the term continuous current is used herein to describethis mode. FIG. 43A illustrates the voltage waveform for the continuouscurrent mode at terminal S in FIG. 29. FIG. FIG. 43A illustrates thevoltage waveform resulting from parasitic effects in the devices. Itwill be observed that from the I_(max) point as transistor FET 50 isswitching off, that the transition time, indicated in FIG. 43A by thedashed line area to its S low voltage state is not zero. It takes acertain amount of time for FET 50 to turn off. The amount of time isbased on the physics of FET 50 and various parasitic capacitanceeffects. Prior to the time that the voltage at terminal S istransitioning from high to low, it will be appreciated that at the highstate when FET 50 is fully on, there is I_(max) current flowing throughFET 50, but of course since FET 50 is fully on, there is no voltagedrop, therefore there is no appreciable power dissipated by FET 50. Atthe other extreme, I_(max) may still be flowing, but it's flowing at adifferent circuit because FET 51 transistor is completely switched off.Since no current is flowing through FET 50 even though there is avoltage across it, no power is dissipated, therefore the power that isbeing dissipated by FET 50 is actually occurring during the time whenFET 50 is turning off, at which point I_(max) is continuing to flow andthe voltage across transistor FET 50 is declining in more or less astraight line fashion towards its off state. Thus power is going to bedissipated in FET 50 and it's wasted because it is not delivered to theload. It will be noted that at a point after FET 50 is switched off, thevoltage waveform continues to fall below zero volts. Inductor L50 isattempting to continue to conduct at I_(max), but neither transistor FET50 nor FET 51 are on at this point, so the voltage continues to falluntil it reaches the switching threshold of diode D50, at which timediode D50 conducts and I_(max) flows through it. If nothing were done,then for the entire time that the S signal is low, current would beflowing in diode D50 with its intrinsic voltage drop. These currents canbe quite substantial (many amperes) so the power dissipated in thisexample can be considerable if allowed to continue for a significantportion of a switching cycle. For a diode voltage drop of 0.6V a currentof 2A and a duty cycle of 50% some 600 mW would be dissipated by thediode. If the power supply were designed to produce 3V at 2A then 10% ofthe power would be consumed by the diode. A synchronous rectifier, wherethe voltage drop of the diode is reduced for most of the time of eachcycle is implemented by FET 51. By reducing the voltage drop acrossdiode D50 and conducting the current at a lower voltage, less power iswasted. As illustrated in FIG. 45C, FET 51 (indicated in the figure bygate drive waveform LF) is not turned on coincident with FET 50 turningoff (indicated in the figure by gate drive waveform UF), because at sometime during that falling waveform, both field effect transistors wouldbe on, thus conducting excessive current and wasting even more powerthan we were attempting to preserve. Therefore, there is a delay, thatis FET 50 must switch completely off, the signal to its gate to turn offis asserted, then sufficient time must elapse for it to actually switchoff. Then the gate control signal to FET 51 is switched on, and then italso has a lag time in responding until FET 51 is completely switchedon. At the right hand side of the waveform in FIG. 43A, the S voltagewaveform is in the low state indicated in the figure by I_(min). At thispoint, FET 51 turns off, and again there is current in diode D50. Thiscurrent will be lower, in fact it's at the I_(min) point on the constantcurrent diagram in FIG. 43, however still enough voltage for the diodeD50 to conduct and the transistor FET 50 is turned on. This lag timeagain is necessary in order to prevent the possibility of bothtransistors being on simultaneously. When FET 50 is turned on, we havethe same situation we had at the falling edge; as the voltage across theFET 50 rises, indicating that FET 50 is turning on, there is a time atwhich current is flowing in upper FET 50, that is I_(min), when FET 50has not yet been fully turned on, and therefore power is dissipated inthat FET 50 and not delivered to the load. There is power loss thereforeat both ends of the switching cycle; however, obviously more power islost at the I_(max) point than at the I_(min) point. The foregoingrepresents a conventional waveform for a synchronous buck switchingpower supply, synchronous meaning that both FETs are turned on during atleast part of the cycle.

Discontinuous Operation

Assume that in circuit 49 of FIG. 29, our load current was for example 2amps, and our period was 2 microseconds. If we drew a map of what theripple would look like, we get 10 volts in and 5 volts out, so we knowthe duty cycle is going to be approximately 50%. The output voltagebeing given by the approximate equation of input voltage times dutycycle equals the output voltage, duty cycle for the FET 50. Then wewould see a waveform as shown in FIG. 30 in terms of the current ininductor L50, which goes from high 2.1 amps down to a low of 1.9 amps,yielding an average current of 2 amps. Capacitor C50 is there tointegrate this current and that is really what you see. This is the socalled “continuous mode”. In this example of continuous mode, FET 50 andFET 51 are each on for one-half of the 2 usec time period. Since theload current is greater than ½ the ripple current, current is alwaysflowing in the inductor and is never allowed to go to zero. Note thatthe current always goes in the same direction, but reverses its slope.Sometimes it's increasing, sometimes it's decreasing, but the currentnever goes to zero. Now, consider that same circuit used to supplyexactly 100 milliamps. As illustrated in FIG. 31, we would have the sameripple and the same 2 microsecond period, the current would go from ahigh +200 milliamps down to 0, with an average of 100 ma. What if wewant less? Well, there are a couple of things you could do. Let usexamine one simple example, suppose we want zero. If we were to run FET50 and FET 51 out of a phase with each other, then if we wanted 0milliampers, and because alternately we have 5 volts supply across coilL50 when FET 50 is on (10v-5v), and 5 volts supply across the coil whenthe FET 51 is on (0v-5v), its ripple current still must be 200 milliampsso how can you get 0? Well, you get 0 because the ripple would look likethat illustrated in FIG. 32. There is −100 ma at time 0 and at 2 usec.Then +100 ma at 1 usec resulting in the current delivered to the load is0. In order for the current to be negative in inductor L50, it meansthat at the points where the current crosses the time axis, the currentactually reversed direction in inductor L50. FET 51 is on for exampleduring this phase and inductor L50 was completely discharged, thencharged in the opposite direction. Now, the current flows in the otherdirection, the negative direction. It's flowing out of the load insteadof into the load. When we turn off FET 51, inductor L50 flies positive.Current flows first through the intrinsic diode in FET 50 and the nowclosed FET 50 closes, actually sending energy back into the power supplyfrom before, and around and around it goes. This is a really poor way tomake 0 milliamps because 200 milliamps are being shipped through all ofthe parasitic losses in the circuit two ways to make no current. Theright answer for making no current is not to do anything, leave bothFETs off. Discontinuous mode, which is illustrated in FIG. 33, is moreefficient at low currents. Using the same duty cycle, and slope, we turnoff FET 50 and turn on FET 51 when the current gets to . . . forexample, 100 milliamps, which is 25% of the cycle time, then the currentramps down to 0 by 50% of the cycle time. This means an average of 50 mAof current was flowing for the first 50% of the cycle. During the secondhalf of the cycle, 0 mA is flowing in the coil and with both FETs offzero volts are across the inductor and 0 mA continues to flow for thesecond 50% of the cycle. Considering these percentages, what results is25 mA over the full 2 microseconds, the total switching cycle in ourexample. The current was not continuously flowing in inductor L50, hencethe term discontinuous current. Between 50% and 100% of the 2 usec timeperiod both FET 50 and FET 51 are turned off.

If we need to reduce this current still further, then the pulses drivingthe gates of FET 50 and FET 51 would be made shorter and shorter untilat some point because of the dynamic that turns the transistors on andoff, the pulse required would become so short that it would beimpractical to make. In other words, the time of the pulse would beentirely consumed by the rise and fall time of the FET. In thiscondition, you actually put a pulse out and wait a number of cycles andthen put another pulse out. This is called cycle skipping which is justa more extreme case of the discontinuous mode of operation. Cycleskipping is illustrated in waveform C of FIG. 43.

As an example of how much of a problem this is, consider power suppliesof the present invention which are designed to provide the standby powerfor a PDA which is asleep. The current consumed by the SDRAMS in the PDAwhen it's asleep is on the order of 2 ma. With a 10 uf filter capacitorsuch as C50, it would take 150 microseconds for the voltage to decay the30 millivolts that may be typically used in the dead band of a controlloop. That is 75, 2 usec cycles skipped between each pulse used to drivethe FETS. For some power supplies in this mode only 150 uA may beconsumed. Under these conditions many seconds may elapse betweenindividual pulses.

In continuous current mode the steady state duty cycle changes a verylimited amount as the current is changed. Once the current falls below ½the ripple current, the desire to maintain efficiency suggests that thecurrent be allowed to become discontinuous. In discontinuous mode, thecurrent delivered is scaled by the amount of time both FETs are off andtherefore the timing of the pulses delivered to the FETs must changerapidly with current. A different regulation algorithm may be useddepending on which mode is required by the load current. Mode changescan represent a significant challenge in prior art implementations basedon analog comparators and amplifiers. On a digital basis, the circuitryof the present invention, can detect that the load current is less than½ our calculated ripple and simply solve for the correct FET timingbased on discontinuous mode. This change effects only the calculationbut has no effect on the structures utilized for the output drivers, theA to Ds, or the sample and hold circuitry. Accordingly, during thediscontinuous operation, we don't have the inductor feeding current backto our supply. So we avoid the so-called negative current, negativecurrent meaning the current flowing from the coil back out to the powersupply. In discontinuous mode, a major objective is to prevent thecurrent in the coil from becoming negative. In this example the outputvoltage is ½ the input voltage implying that FET51 would need to be onfor exactly the same length of time as FET50. There is more subtletythan this because of the intrinsic diode D50, which exists across FET 51from terminal “S”, which is the point where the inductor L50 intersectsFETs 50 and 51. The purpose of the synchronous transistor FET 51 is toreduce the voltage drop across the FET when there is current flowing.There is a hazard in waiting too long to turn off FET 51 because thiswould cause the current in the diode to be reversed creating a largeringing waveform at terminal S when FET 51 eventually opens. Thisringing is undesirable because of the interference it produces as wellas a small negative impact on efficiency. One way to mitigate thispotential problem is to turn FET 51 off slightly before the point atwhich the current in inductor L50 would cross 0 current. Since thecurrent is quite low at this point, there is very little penalty interms of power In our implementation, we utilize algorithms inregulation control module (REG) 1204 to control theconduction/non-conduction of the FETs. In the prior art, the current inthe inductor is measured continuously and an attempt is made to detectthe instant where the current in the inductor crosses 0, and thenrelease the drive to the inductor. A problem that is encountered is thatthe zero current point is often detected late because there is apropagation delay time in the comparator and on the switchingtransistor. In an attempt to solve this problem, “a ring killer circuit”may be used, which is another transistor placed across the inductor,after the lower transistor is off, the transistor is turned on acrossthe coil to discharge it.

Reference to FIG. 43B and FIG. 43C in connection with the followingdescription illustrates that the synchronous mode of operation is notdesirable in some load situations. First consider FIG. 43B which shows avery long duty cycle, that is the time that the S signal is low(indicated by A in the figure) is very short. This can occur when theinput voltage is very close to the output voltage. Coincidentally, thisis also a time when switching power supplies may be their most efficientat least in the buck converter configuration. If we assume, for example,that time A is equal to 100 ns, there is not sufficient time to turn onFET 51, and if one tried to do that, it would have to be turned offalmost immediately risking an overlap with the conduction of FET 50.Thus for a very long duty cycle, that is FET 50 conducting for a highpercentage of the total cycle, it is not desirable to use FET 51. Thislimitation will typically define the maximum duty cycle that asynchronous buck switching power supply can attain, that is it canattain a duty cycle that is so long that there is insufficient time toturn FET 51 on and off. In accordance with one aspect of the presentinvention, the duty cycle of FET 50 is monitored by regulation controlmodule (REG) 1204, and if the duty cycle is sufficiently long, thendynamically the system is configured such that no gate drive signal isapplied to FET 51 so that is never conducts. This dynamic operation isdescribed fully below.

FIG. 43C illustrates a plot of the voltage at S as a function of time ina mode of operation that is the other extreme of operation, that is atextremely low load currents. The S waveform of this figure correspondsto the discontinuous current mode such as that illustrated in waveform Bof FIG. 43. The initial voltage at terminal S is between the inputvoltage V_(in) and ground, that is the initial voltage would be at theoutput voltage V₀; that is if the current in inductor L50 is zero, thenthe voltage across inductor L50 must also be zero. From FIG. 43C, itwill be noted that transistor FET 50 is switching on before generating ashort on pulse. The duty cycle of FET 50 is indicated in FIG. 43C and itwill be appreciated that FET 50 is on for a very short time. When FET 50is off, inductor L50 forces the S term down to ground and then belowground, in which case diode D50 conducts as before. Then the current indiode D50 decays as the very small amount of current in inductor L50decays until eventually the current in inductor L50 is no longersufficient for diode D50 to conduct. The voltage on inductor L50 risesagainst the various parasitics, that is the voltage at terminal S risesagainst the various parasitic capacitances in the circuit until again itreaches the output voltage V₀ where it remains until the next pulse isdelivered, that is when FET 50 is turned on at the beginning of the nextcycle. In this mode, it is not advisable to use FET 51, not because ofinsufficient switching time to turn FET 51 on, but because there isinsufficient current to sustain inductor L50 below ground for longenough for FET 51 to be turned on. An attempt to turn FET 51 on wouldlikely result in it being on too long, and the current in inductor L50would actually reverse direction, that is flow from the load to ground,resulting in a ring effect when FET 51 is turned off. Thus in this mode,to avoid the inefficiency resulting from current flowing out of theload, and also the interference caused by this ring effect, it isdesirable to operate in a non-synchronous mode in which FET 51 would notbe turned on during an operational cycle. Thus for extremely short dutycycles, the system dynamically changes from synchronous tonon-synchronous operation based on the duty cycle of FET 50 fallingbelow a predetermined minimum value. It should also be noted that forlonger duty cycles in a discontinuous current mode, FET 51 should beturned on for a part of the cycle, but not the entire cycle. Theabove-described discontinuous current in waveform C in FIG. 43 forterminal S is one that would be typical of what would be seen in cycleskipping. In the case of discontinuous current where a pulse isgenerated on every cycle as shown in waveform B of FIG. 43, there may bea time long enough that the FET 51 could be turned on for a time whichis calculated to be less than the time required to discharge inductorL50, and then FET 51 would be turned off so the inductor L50 cancomplete its discharge through diode D50, which would automatically turnoff when its current became zero thus avoiding the ring effect.

Switching power supply controller 1200 calculates when a power supplycircuit such as circuit 49 should be changed from a synchronous tononsynchronous mode based on a number of factors. For example, assumethat FET50 is required to be on for a relatively long portion of thecycle time in order to produce V₀ having a magnitude near the magnitudeof V_(in). In one mode of operation, the manufacturer's operationalcharacteristics of FET51 are stored in nonvolatile memory 1216. Ofparticular relevance to this operation is the turn-on/turn-off time forFET51. As noted above, if the portion of the duty cycle that FET50 mustbe on is relatively large, then the time available to turn FET51 on andthen off before the next cycle may not be sufficient. Regulation controlmodule (REG) 1204 in conjunction with system hardware central processingmodule (SYS) 1205 perform a computation, based on the pulse width neededto drive FET50 and the remaining time in a cycle for a pulse to turnFET51 on and then off, of whether a gate pulse should be generated toturn FET51 on and then off during the remaining portion of a cycle. Inaddition to the required on-off transition time of FET51, thepropagation delay in driving the gate of FET51 is also considered in thecomputation. If the time remaining in a cycle is insufficient to turnFET51 on and off, then regulation control module (REG) 1204 does notsend a command to digital pulse control wrapper 1201 to generate a drivepulse to the gate of FET51, thus providing dynamic change fromsynchronous to nonsynchronous operation of the power supply beingregulated by switching power supply controller 1200. In an alternatemode of determining whether FET51 should be turned on at all in a cycle,switching power supply controller 1200 examines the power lossconsideration. Even if the time remaining in a cycle is sufficient toturn FET51 on and off before the beginning of the next cycle, from apower loss standpoint it may be preferable not to do so. For example, ifthe system calculates based on the current flowing in inductor L50 thatthe power consumed by the conduction of FET51 will be greater than thepower that would be consumed by intrinsic diode D50, then no gate driveis applied to FET51 and the regulated power supply is operated in thenonsynchronous mode.

At the other extreme, where FET50 needs to be on for a relatively shortportion of an operational cycle, then it may not be appropriate to turnFET51 on. The system calculates, based on the current through inductorL50, the amount of time it would take for the current in L50 to drop tozero after FET50 is turned off. This computation is possible because thecharacteristics of FET51 and the other components of supply 49 arestored in NVM1216. If the computed time for the current to fall to zerois less than the time it would take to turn FET51 on and off, it ispreferable not to utilize FET51. Instead, the current is permitted todecay to zero through intrinsic diode D50. In an alternative mode, thepower consumption can be computed to determine the advisability ofturning FET51 on and off. Based on the results of a comparison of powerconsumed if FET51 is used and letting the current decay through diodeD50, the system determines if the operation should be synchronous ornonsynchronous.

The above analysis considers the synchronous versus nonsynchronousoperation of a buck circuit. Similar analysis of the operation and thedecision to operate it in a synchronous or nonsynchronous mode is madeby switching power supply controller 1200 for a boost circuit or SEPIC.Consider the circuit shown in FIG. 46, and assume that QT_(Ø), QB_(Ø),LØ, RØ and CØ are being controlled as a boost circuit by switching powersupply controller 1200.

Further assume that the Batt. Ø has an output voltage which is slightlybelow that needed for Supply A. Switching power supply controller 1200,based on the voltage of Supply A, sensed over lead F1, and the batteryvoltage sensed over lead S2, configures the circuit to a boostconfiguration based on the sequence of gate drive pulses to the gates ofQT_(Ø) and QB_(Ø). Because the difference between the battery voltageand the required voltage of Supply A is small, the gate pulse providedto QB_(Ø) will have a relatively short duration. If the circuit were tobe operated in a synchronous mode, after QB_(Ø). was turned off, QT_(Ø)would be turned on to deliver current, and hence power, to chargecapacitor C_(E) to the slightly higher voltage required. The magnitudeof the current through RØ is computed by switching power supplycontroller 1200 since the resistance of RØ is known to the system,having been prestored in nonvolatile memory 1216. Similarly, thecharacteristics of the transistors are also prestored in non-volatilememory 1216. Knowing the foregoing parameters, switching power supplycontroller 1200 computes and compares the power losses for the scenarios(i) in which QT_(Ø) is turned on and (ii) where QT_(Ø) is not turned on.As will be recognized by those skilled in the art, even if QT_(Ø) is notturned on, current will flow through it because of the intrinsic diode(not shown) of QT_(Ø). If the result of this computation and comparisonshows that less power will be consumed by not turning on QT_(Ø), thenswitching power supply controller 1200 will not provide a gate pulse toQT_(Ø), resulting in nonsynchronous operation. It will, of course, beappreciated from the above that the decision to operate eithersynchronously or nonsynchronously is made dynamically by the system,unlike the prior art in which typically the user sets the circuit tomode of operation to one or the other, and any change must be mademanually by the user.

It will be appreciated that in above we have described an architecturewhere a synchronous rectifier is used in a switching power supply onlyfor the operating regimes where it improves efficiency, while in otherregimes of very long and very short duty cycles, the synchronousswitching feature is dynamically eliminated to avoid inefficiency.

Section 1.2.2.12 Digital Resonance Control Loop

FIG. 44 illustrates a half-bridge high voltage power supply circuit,indicated by reference character 1.2.2.12, which can be used to power acold cathode fluorescent light bulb CCFL1. In a half-bridgearchitecture, energy is applied first to one side, then to the other.Inductors L12 and L13 replace the upper two transistors that what wouldcomprise a full bridge. Half-bridge circuit 1.2.2.12 illustrated in theabove-named figure includes in a first leg inductor L13 which isconnected in series with FET A between a 5V supply and ground.Similarly, in the second leg, inductor L12 is connected in series withFET B between the 5V supply and ground. The gates of FET A and FET B aredriven, for example, with waveforms WF1 and WF2 respectively or WF3 andWF4 are respectively shown in FIG. 44A. The drive control signals forthe gates of these transistors are provided, for example, by NFETdrivers module 1202 illustrated in FIG. 12. Feedback signals from senseresistor R12 are provided over lines C1 and CØ to the sample and holdmodule 1207 also illustrated in FIG. 12. The duty cycle of gate drivesignals are adjusted as a function of the changes required to provide anappropriate voltage for CCFL1. Piezoelectric transformer PZT1 may beimplemented, for example using a transformer KPN 6003A from CTS WirelessComponents located at 4800 Alameda Blvd., NE Albuquerque, N.Mex. Similardevices may be, of course, substituted. As illustrated in FIG. 44, theinput to terminal T1 of PZT1 is provided from the common connectionbetween inductor L13 and the drain of FET A, and the input to terminalT2 is provided by the common connection between inductor L12 and thedrain of FET B. The output of piezoelectric transformer PZT1 is coupledto one side of CCFL1. The other terminal of CCFL1 is connected to theupper terminal of sense resistor R12, the lower terminal of which iscoupled to ground.

Although circuit 1.2.2.12 is driving a piezoelectric transformer, itcould be used to drive a conventional magnetic transformer.Piezoelectric transformers differ from magnetic transformers in that themethod of converting from a lower voltage to a higher one, with aproportionate decrease in current, is electromechanical in nature, butotherwise they have similar characteristics, that is, they come involtage in to voltage out ratios. In one embodiment of the presentinvention, PZT1 has a mechanical advantage of 100 to 1. This means thatfor every volt impressed across the input terminals T1/T2, 100 voltswould be provided at output terminal OT. Piezoelectric transformers,since they are electromechanical in nature, have a natural resonantfrequency and they will not tend to operate outside of a relativelynarrow band of frequencies that are determined by the mechanicalcharacteristics of the device. This frequency can vary from device todevice. That is, it's not completely process controlled in themanufacture, and although the device specification provides a value forthe expected resonant frequency, the actual resonant frequency can varymany percentage points one way or the other. The optimal efficiency isobtained at the resonant frequency and operation far enough away fromthe resonant frequency will actually cause the device to fail tooscillate. FIG. 44A shows several examples of driving waveforms.Waveforms WF1 and WF2 are applied to the gates of transistors FET A andFET B, respectively. These waveforms show the maximum drive amplitudethat would be possible which would be each of these waveforms at 50%duty cycle, 180° out-of-phase. A cycle for WF1 is indicated in FIG. 44A.A cycle for WF2 is the same length of time; however the start time isoffset. This would result in a driving waveform of approximately 5 voltsto piezoelectric transformer PZT1 and when operated within resonance,this would result in approximately 500 volts AC being applied to CCFL1.As will be appreciated by reference to FIG. 44A, waveforms WF3, WF4 havethe same frequency as waveforms WF1 and WF2, however they have shorterduty cycles. Applying these shorter duty cycles reduces the energydelivered to PZT1 and correspondingly provides essentially a function ofcontrolling the voltage and current output of the piezoelectrictransformer. With this circuit, it is important to discover the resonantfrequency of PZT1 and remain on the resonant frequency. Severalalgorithms are possible. One example of a resonance algorithm is tochange the frequency of the drive signals to terminals T1 and T2 whileobserving the feedback signals C1 and CØ. At frequencies which are faraway from the resonant frequency, there would be no feedback seen, as novoltage should be applied across the CCFL1. Starting from a frequencybelow the minimum listed by the manufacturer of PZT1, as the frequencyof the input drives to the gates of FET A and FET B is increased, CCFL1will begin to ignite, and a signal will be detected at C1, CØ. As thefrequency is increased, the phase relationship between the signal C1, CØand signals WF1, WF2 will begin to shift, and resonance is indicated atthe point where a 90° phase shift is observed. A second way to determinewhen the resonant frequency of PZT1 has been reached is to examine theamplitude of the signal at C1, CØ, knowing that PZT1 has a maximumoutput at its resonant frequency. The frequency of the gate drivesignals is swept until a voltage appears at CØ, C1 indicating that thecurrent is flowing through CCFL1, and then the frequency can be furtherramped and the voltage at C1, CØ monitored until a peak is arrived at,again indicating resonant operation. It should be noted that thetopology could be supported in combination with the other topologiessuch as buck, boost, and SEPIC. When using synchronous sampling, alsosometimes referred to herein as adjacent phase sampling, as thefrequency of drive signals WF1, WF2 is changed, scheduling electronicsin regulation control module (REG) 1204 may need to vary the edgesslightly for WF1, WF2, so that the voltage and current samples beingtaken for the other supplies being controlled by switching power supplycontroller 1200 are not corrupted, and such that the samples taken fromCØ, C1 on this supply will not be corrupted by the gate drive signalsfor the other supplies.

Section 1.2.2.13 Linear or Fold Back Current Limiting from the SameStructure

Power supplies can be designed to regulate voltage, current or power.Constant voltage power supplies are used for things like microprocessorsand memory devices and other voltage operated devices. In the case of apower supply which is designed to regulate voltage, the power supply mayhave a maximum current which is specified for safety or for noisereasons or other reasons and the power supply will regulate in voltageuntil such time as the current exceeds a preset limit. This limit istypically set by external components like resistors or similarcomponents. At that point, a fault condition exists in the power supplyand the power supply will go back to a very low current. This techniqueis known as fold back current limiting. This provides protection in thecase of a short circuit condition presented to the power supply, byhaving the power supply limit the maximum current to a very small numberso that neither the power supply nor whatever element that caused thetemporary short circuit would be damaged. The power supply in the foldback mode does not regulate the current, but rather limits it to a verysmall number, and its main operational mode is in voltage. An alternateapplication for a power supply is in constant current. Constant currentpower supplies power devices that rely on current for their function, anexample of this kind of a device being an LED (light emitting diode)which produces light in proportion to the amount of current that isflowing in it. The voltage applied to the LED is not particularlyimportant and in fact it varies depending on temperature and otherfactors, but the current will always produce a similar proportionalamount of light fairly independently of other parameters. A constantcurrent power supply does not examine the voltage in its control loop;it examines the current and attempts to regulate the current through theload to be a constant number, essentially supplying whatever voltage isnecessary to maintain that constant current. It can be seen from thisthat a constant voltage power supply with fold back current limiting hasa control loop which regulates the voltage and watches the voltage,treats an over-current condition as a fault and then takes protectiveaction to make the current very small until the short is removed and thevoltage is allowed to rise. A constant current power supply regulatesthe current and a fault condition for a constant current supply can bethat the voltage has risen to too high a level, thus indicatingtypically that the load has open circuited. In one implementation of theinvention, all of the feedback terms, current and voltage are derivedfrom analog to digital converter analog to digital converter 1206 (FIG.12). Thus the feedback is turned into numbers at this point before anycontrol is attempted to be applied. Similarly the control output pulsewidth modulated signals are also digitally controlled. In between theseis a processing element in regulation control module (REG) 1204, whichcan run a variety of algorithms for any channel controlling a powersupply. The algorithms can, for example, regulate voltage producing aconstant voltage power supply with fold back current limitingcharacteristics for error, or regulate for constant current whereexcessive voltage would be a fault, all without altering the samplingstructures such as sample and hold SHM 1207 (FIG. 12), analog to digitalconverter 1206 (FIG. 12), or the DPC 1201 (FIG. 12). An example of anapplication for this general purpose capability is in battery charging.For a battery using lithium ion chemistry, a constant current should beprovided for the first part of the charge cycle, and then a constantvoltage should be applied until the current has fallen below a minimumlevel. In this operation, the same power supply circuit can becontrolled by switching power supply controller 1200 (FIG. 12) and theoperational control mode for the power supply circuit changed fromconstant current to constant voltage. The hardware for implementing theforegoing includes regulation control module (REG) 1204, the SHM 1207,analog to digital converter 1206, and DPC 1201. This hardware and thecontrol loops are described in other portions of the application.

Section 1.3 Converter Topology/Topology Equivalents Supported

Section 1.3.1 Buck, Boost, Sepic, Sync, Half-bridge, Multi-phase etc.from a Single Controller Structure

The buck converter illustrated in FIG. 45 derives its output voltage,neglecting parasitic effects, by the following straight forwardequation:V _(o) =V _(in) ·DC _(UB)

Where:

-   -   V_(o) is the output voltage    -   V_(in) is the input voltage, and    -   DC_(UB) is the duty cycle of transistor UB

In the above equation, duty cycle (DC) is the percentage of the totalcycle that transistor UB is on expressed as a decimal number. Forexample, if the duty cycle (DC) is 50%, then V_(o) is one-half ofV_(in). FIG. 45A illustrates the current I and output voltage V_(o) as afunction of the conduction of transistors UB and LB in FIG. 45. As willbe noted in that figure, the voltage and current cycle is indicated byCycle in the figure and spans the time elapsed between the leading edgesof the gate drive signal to transistor UB. This means that there's adirect proportional relationship between the duty cycle in the upper FET(UB) with the output voltage and the duty cycle is the proportion ofinput voltage to output voltage. So this means that if the outputvoltage were to fall below a preset value that a small increase in dutycycle could correct the error. If the voltage were to be higher than itshould be, then a small reduction in duty cycle would put the voltageback where it was supposed to be.

FIG. 45B illustrates boost supply 1.3.1B. In the boost supply, theaction is quite different. More particularly, as pointed out above,power is sent to the inductor L1.3 by transistor UB in the buck supplyof FIG. 45. In contrast, in the boost supply illustrated in FIG. 45B,the conduction of transistor LF energizes the inductor L1.3B. Referringto the timing diagrams in FIG. 45C, it will be appreciated that thecurrent ramp in inductor L1.3B is started by the transistor LF, where asin the buck converter of FIG. 45, the current ramp is started by theupper transistor UB. The output voltage V_(o) may be expressed by thefollowing formula:$V_{o} = {V_{i\quad n} \cdot \frac{1}{1 - {D\quad C_{LF}}}}$

Where DC_(LF) is the duty cycle of transistor LF.

In the buck converter of FIG. 45, a fifty percent duty cycle results inan output voltage V_(o) that's one half the input voltage V_(in). In theboost converter of FIG. 45B, a fifty percent duty cycle results in anoutput voltage V_(o) that is twice the input voltage V_(in). Inaddition, as the duty cycle of transistor LF increases, the outputvoltage V_(o) increases for the boost supply. For the buck supply, asthe duty cycle of transistor UB increases, the output voltage V_(o)increases. So it will be seen from the above that if one were toconstruct a feedback loop for the buck supply, wherein a change in theoutput voltage would result in a proportional change in the oppositedirection for transistor UB, if one then took that same feedback networkand applied it to the boost supply, the control would be reversed. Thatis, for the boost circuit (FIG. 45B), an increase in the output voltageis desired, and the duty cycle of UF was increased, thereby reducing theduty cycle of transistor LF, output voltage V_(o) would actually go thewrong way, so the control loop would have to be reversed. In addition tothat, if one were to attempt to use the absolute values of the dutycycles, that would not work because of the inverse proportionality inthe case of the boost supply, so it should be clear that these twoswitching power supplies need a reversed sense of feedback and they needdifferent feedback mechanisms and a different sequence of control. As anexample, in the buck converter, it will be noted from FIG. 45A thattransistor LB turns off before transistor UB turns on during a powercycle. But in the boost circuit (FIG. 45B), the transistor LF turns onafter transistor UF turns off in a power cycle. So, again, the sequenceof events, which signal provides the power, which signal is the leadingsignal, is actually reversed.

Switching power supply controller 1200 (FIG. 12) is adapted to alsocontrol what is referred to in the art as a single-endedprimary-inductance-converter (SEPIC) circuit. A typical SEPIC circuit isillustrated in FIG. 46, and indicated by reference character 1301.3. Inthis circuit, Batt. 3.3, which provides Supply B voltage, drives one endof the primary (indicated by reference character P) of transformer T3.3.The other end of the primary is connected to FET 3.3. Input capacitorC3.4 is connected across Batt. 3.3. Capacitor C3.3 is connected betweenone terminal of FET 3.3 and one terminal of the secondary (indicated byreference character S) at transformer T3.3 as well as to one terminal atFET 3.4. Transistor FET 3.4 is coupled between one terminal of thesecondary S and one terminal of sense resistor R3.3, the other terminalof which provides V_(out). Although intrinsic diodes are onlyillustrated across FETs QB₁, 3.4 and 3.3, they are inherent in all ofthe field effect transistors. Capacitor C3.4 is connected between theV_(out) terminal and the common ground for the circuit. Feedbacks toswitching power supply controller 1200 are provided over lines S6, S7,S8 and S9. The gate drive signals to FET 3.3 and FET 3.4 are provided byswitching power supply controller 1200. The operation of the SEPICcircuit will not be described here since it is well known to thoseskilled in the art. However, with regard to the switching signals to thegates of the transistors, it will be recalled that in a first phase FET3.3 conducts and in a second phase FET 3.3 is turned off and FET 3.4conducts. The timing and duration of the control signals to the gates ofthe transistors is determined by the signals from switching power supplycontroller 1200. Here again, as in the above-described buck and boostcircuits, the order of and duration of the gate control signals iscontrolled by switching power supply controller 1200 to provide thedesired output voltage V_(out) at a target constant value oralternatively at a constant current. The operation of circuit 1301.3 is,as with the other circuits which are controlled by switching powersupply controller 1200, is determined by a system configuration which isdescribed elsewhere herein. The configuration is programmed at theoutset of the operation of the power supply circuits connected toswitching power supply controller 1200. As described in more detailbelow, the operation of the circuits can be changed dynamically byswitching power supply controller 1200 as a function of the feedbackfrom the power supply circuit being controlled, the application ofexternal power, and among other things the voltage of batteries used inconnection with the circuit.

Switching power supply controller 1200 may also support other powersupply configurations such as the half-bridge which is illustrated inFIG. 25 and FIG. 44. As explained more fully below, regulation controlmodule (REG) 1204 includes transfer functions and is programmed toprovide the appropriate signals for controlling power supplies connectedto the system. In one embodiment, DPC 1201 and NFET driver module 1202are utilized to generate the control signals which are provided to thegates of the power supplies connected to the system.

In switching power supply controller 1200 (FIG. 12), because thesampling function performed by SHM 1207 and the driving functions aremerely controlled by analog to digital converters in the case ofsampling functions, and digital logic in the case of the drivefunctions, it's a software task, or one method of implementation is tosimply apply the correct transfer function and sequence of control inorder to change topology. In this way, with the same internalstructures, drivers for the external FETs, sample and holds andmultiplexers, analog to digital converters for reading the inputvoltage, a variety of control loops supporting a variety of differentexternal topologies can be implemented without change in the external orinternal structure of the chip except with regard to the software thatruns within it.

We've demonstrated above three different topologies that are verydifferent, but it should be seen from this that actually any number ofdifferent topologies could be implemented so long as their transferfunctions are known and the circuit itself has been informed of theexternal topology and the interconnect of the external components.

Section 1.3.2 Buck, Boost, Sepic, Sync, Half-bridge, Multi-phase etc.from a Single Controller Structure

Referring to FIG. 12, it can be seen that in this implementation, thereis a single regulation control module (REG) 1204 which controls avariety of outputs. By providing programmable information to thesoftware data set for each instance of the regulation softwareassociated with each output, different topologies (i.e., buck, boost)can be achieved on different outputs simultaneously. For example, oneset of outputs might be described to the regulation hardware block asbeing connected in the configuration of a buck converter. An adjacentset of outputs could be defined to the regulation hardware block as aboost converter or a half bridge, a sepic, or other topology. Regulationcontrol module (REG) 1204 then can dynamically by processing first onefeedback and then the next, switch between different topologies alloperating on different pads from the point of view of the externalsystem all operating simultaneously, but due to the sample nature of thedata and the digital nature of the pulse width modulation control, infact, a single regulation engine is processing each topology andfeedback strategy in time one after the other to maintain regulation onall pads simultaneously.

Section 1.3.3 on the Fly Topology Reconfiguration

In one implementation of the present invention, dynamic reconfigurationof a controlled power-supply is provided. In one mode, the controlledpower supply may operate as a buck supply, in another mode as a batterycharger and in another a battery boost circuit. Referring to FIG. 46, anembodiment is illustrated in which the topology of circuit 1300.2 can bechanged by appropriate application of control signals to the gates ofthe transistors. For example, assume that no external power is beingsupplied and that Batt. Ø is at 3.1 volts. Further assume that circuit1300.2 is running as a boost converter off of Batt. Ø and providing 3.5volts to Supply A, which is driving Load I through Supply 1. In thisexample, we further assume that Batt. Ø when fully charged has an outputof 4.2 volts. If external power (indicated in the figure by Ext Pwr) isapplied, of for example 12 volts, this is detected by switching powersupply controller 1200. Upon detecting that external power is present,switching power supply controller 1200 provides drive control signals tothe gates of transistors QT_(Ø) and QB_(Ø) to operate circuit 1300.2 asa buck power supply, allowing current to flow from the external powerand be regulated through circuit 1300.2 and then delivered to Batt. Ø atthe correct current and voltage to affect charging of Batt. Ø. If Batt.Ø becomes fully charged, then the circuit 1300.2 could be switched off,or maintained in a trickle charge mode to maintain the charge on Batt.Ø. The external power also supplies power to circuit 1301.2, which forillustration purposes, is supplying Load I at 3.3 volts via the buscalled Supply A. In another mode, assume that the external power isremoved and that Batt. Ø is either fully charged to 4.2 volts or chargedto a voltage sufficiently high to drive circuit 1301.2 directly fromBatt. Ø. Switching power supply controller 1200 detects that theexternal power has been removed by observing the drop in voltage ofSupply A. Under these conditions, switching power supply controller 1200will turn transistor QT_(Ø) on continuously and transistor QB_(Ø) willnot be provided with gate drive signals. Switching power supplycontroller 1200 operates in this mode because it detects that Batt. Ø isproviding an output voltage of 4.2 volts or a lower voltage which issufficient to supply voltage to circuit 1301.2 in excess of the 3.5volts which it needs to maintain regulation output to Load I at 3.3volts. This state is maintained as the battery is discharged throughtransistor QT_(Ø), through Supply A into circuit 1301.2. When switchingpower supply controller 1200 detects that Supply A has declined to 3.5volts, which would represent a duty cycle of about 95% for circuit1301.2, at this point it's clear that without additional voltage atSupply A, it would not be possible to maintain load regulation for V_(o)being provided by circuit 1301.2. Accordingly, switching power supplycontroller 1200 transitions circuit 1300.2 to a third state, that is thestate where the battery is less than 3.5 volts and switching powersupply controller 1200 begins to operate circuit 1300.2 as a boostconverter, whose source is Batt. Ø and switching power supply controller1200 regulates boost operation to regulate Supply A at 3.5 volts.Switching power supply controller 1200 can now maintain Supply A at 3.5volts meeting the minimum requirements of circuit 1301.2 until thebattery is discharged or until external power is again available. Thisallows the system to provide output voltages which may be higher orlower than the input battery voltage and allows switching power supplycontroller 1200 to utilize circuit 1300.2 in one of three modes: (i) asa buck converter to charge the battery; (ii) as a switch to supplybattery voltage directly to Supply A through QT_(Ø); or (iii) as asynchronous boost converter to allow operation of circuit 1301.2 toprovide output voltages above the voltage available from the Batt. Ø.The hardware for implementing the foregoing is shown in FIG. 12, andcomprises central processing module (SYS) 1205, regulation controlmodule (REG) 1204, sample and hold module SHM 1207, the output of whichfeeds analog to digital converter 1206. Also, NFET drivers module 1202are utilized to drive the gates of the transistors.

Section 1.3.4 Programmable Topology on a Multi-output Controller

In order to provide support for different topologies in prior artsolutions, the same internal structures would actually have to changetheir sense of feedback, the way the pulse width modulation signal wasfed to the external FETs would have to be reversed, dead timesreinterpreted to mean something different that does not overlap, etc.The foregoing discussion on supporting multiple topologies from the samestructure indicates that if the fundamental elements were digital, thatis, the feedback information monitored from the outside were convertedto a digital signal and the control of the pulse width modulator outputswere also done digitally, that all of this could be accomplished insoftware; that is, the single structure could support multipletopologies. In order to make that a practical production product, it isnecessary for such a switching power supply to know at the instant ofactivation what sort of a supply it is supposed to be. This can beaccomplished in a variety of ways. In one implementation, this could beaccomplished with external mode control pins, which could be soldered inone state or the other. A disadvantage to that strategy is that itwouldn't be possible to change the topology dynamically, which may bedesirable. For example, the circuit of FIG. 45 could be a batterycharger circuit, which is a buck topology for the purposes of chargingthe batteries which would be connected to V_(o), but later, underdifferent circumstances, when external power is no longer available andthe batteries are the source of power, that same external connectioncould actually represent a boost topology, where the input and outputvoltage samples are reversed in their significance and the duty cycle isreversed in its significance and new regulation is applied. Moreparticularly, circuit 1.3.1 would be converted to a boost circuit inwhich the battery connected to V_(o) becomes the V_(in) and the V_(in)would become V_(o). In this scenario, the direction of the currentindicated in FIG. 45 would, of course, be reversed. So it's desirable tobe able to change the topology on the fly and this can be accomplishedby providing mode control bits within the switching power supply itselfthat could be stored, for example, in a non-volatile memory 1216 (FIG.12), or could be changed under program control, for example, to go fromthe battery charging operating to boost operation.

Section 1.4 Discussion of Coulombmetric Measurement in a Switching PowerSupply

Section 1.4.1 Cycle-by-cycle Energy Extrapolation from Current/voltagePulse Data

The current invention offers a significant improvement in a systemdesigner's ability to accurately measure, control, and predict theenergy available to the system from it's rechargeable battery orbatteries. Importantly, this is accomplished without additionalcomponents beyond those already provided for the buck/boost regulationsystem described. Many of the component parasitic values, described inconjunction with the regulation algorithms, are also reused. Finally,the current invention is more energy efficient and enables much-improvedaccuracy in determining the remaining battery energy at any givenmoment, which system designers may translate into longer battery life,smaller batteries, lower weight, smaller form factor, lower cost, or acombination of these.

The prior art does not provide a satisfactory method of accuratelymeasuring the remaining energy capacity of a battery. For example,cellular phones typically rely upon measured voltage to indicate stateof charge. However, if a low battery is recharged, even for a briefperiod of time, it will falsely indicate a full charge when removed fromthe charger. Since this is simply a measure of surface charge, notenergy available, the user will experience the equivalent of shortbattery lifetime, perhaps even believing the battery is worn out andneeds replacement.

Using coulombmetrics, one measures the actual energy delivered to andtaken from a battery. Energy available can be influenced by such factorsas the number of past charge/discharge cycles, cell temperature,charge/discharge rates and such. Fortunately, these factors do not haveto be precisely known if one knows how much net energy has been put intothe battery. The approach is to determine how much energy is put intothe battery, hence available for delivery, then measure said deliveryaccurately and alert the system monitor when certain conditions arereached. This is important for personal digital assistants (“PDAs”),notebook computers, and other devices which hold certain information involatile memory devices but can move the data to a nonvolatile media ifthe unreliability or failure of the volatile memory device is imminent.

For example, PDAs which utilize the WIN CE operating system (“OS”) mustfaithfully save the context of all open files and certain systemvariables before a complete shutdown with attendant loss of volatilememory. In fact, failure to save this data before loss of adequate powermay corrupt the OS, such that the product is rendered irrecoverablynonfunctional. For this reason, such products are not designed solelywith user-removable batteries but rather with at least one non-removablebattery for memory keep-alive. The system design cannot just recognizewhen the energy is depleted, but must anticipate depletion far enoughahead of time to allow adequate energy to save off the critical data toa nonvolatile media.

Since determining battery energy from voltage alone is very inaccurate,designers must allow a large margin. This causes the user to experiencea short battery life, since it is when the product ceases operation, notwhen the battery is actually depleted, that the user perceives thebattery to be exhausted.

Another, better approach of the prior art is to use a measuring deviceassociated with the battery. In notebook computers this device is ofteninside the battery case itself. In PDAs it is an additional, costlydevice, external to the battery. As with the current invention, theseschemes measure the power delivered to and removed from the battery, aswell as measure battery temperature to calculate the energy available.However both approaches suffer from a serious shortcoming: the sensingresistor needed to create a voltage drop sufficient to measure duringlow current draw (say, 2 millamperes in standby operation) must berelatively large, which then causes it to consume a significant amountof power during high current operation.

A unique technique utilized in the present invention is to insteadaccurately measure the power delivered from and provided to the powersupply/regulation system. An important aspect of the present inventionis to indirectly measure current draw during low power operation withoutusing a sense resistor. This resistor, which is needed for regulation,may then be a low value such that there is little loss during high poweroperation.

Referring to FIG. 46, one topology is illustrated showing a single,multipurpose stage 1300.2 which can be used as battery charger stage,and a single output stage 1301.2. In this configuration, assume no poweris externally supplied. The voltage of battery Batt. Ø may range from4.2 to 2.7 volts. If, for example, the target value for V₀ is 3.3 voltsand we connect the Supply A rail directly to a 2.7 volt battery, a buckconverter will not work. In the present invention, when the batteryvoltage is 4.2 volts, we turn on FET QT_(Ø) and leave it oncontinuously, which supplies 4.2 volts to Supply A. Stage 1301.2 thenoperates as a buck converter. When the voltage of the battery approachesthe required output voltage V₀ of the buck converter, we begin operatingstage 1300.2 as a boost converter. More specifically we turn ontransistor QB_(Ø), causing inductor LØ to charge, then dischargeinductor LØ into capacitor C_(E) via transistor QT_(Ø) with transistorQB_(Ø) turned off. This allows operation all the way down to a batteryvoltage of 2.7 volts while maintaining a regulated 3.3 volts at V₀. Thetypical battery voltages will depend upon the technology of the batteryselected for the system.

There are two distinct techniques employed by the invention to measurepower removed from the battery. One is appropriate for very low currentdrain conditions. During low current discharge and when circuit 1300.2is operated as a buck circuit, the time between gate pulses totransistor QT_(Ø) is typically very long. This is typical ofapplications in low power, wherein power is only needed due to leakageand supervisory circuits; usually a few milliamps or less. Referring toFIG. 46B, the higher current method is used when the inductor L3 isdriven during some portion of each cycle time. Note that continuous anddiscontinuous modes (defined elsewhere) utilize the same methodology.All of these techniques have the benefit of utilizing only thecomponents already employed for accomplishing regulation.

First let's examine the measurement of the power taken from the batteryduring low current operation. One can monitor the power delivered by thebattery Batt. Ø by measuring the voltage drop across sense resistor RØ.As discussed earlier, however, this would have the same problem as theprior art, namely the necessity of a large resistance to enable anadequate voltage drop for analog-to-digital conversion (ADC), whichwould then cause a significant power loss during high current operation.The present invention avoids this issue entirely by instead monitoringthe voltage across the Supply A capacitor CE. Since switching powersupply controller 1200, FIG. 25) has an accurate time base from thecrystal oscillator, the power may be precisely determined by the formulaI=C(dV/dt), where C is the value of capacitor C_(E). The change inSupply A voltage (measured by analog to digital converter 1206 shown inFIG. 12 relative to ground) is determined by measuring and recording thevoltage at precisely known time intervals, which are under programcontrol. The advantages of this method are numerous, for example: (i)the value of resistor RØ, may be kept very small; (ii) all of the powerremoved from the battery is measured; and (iii) no additional componentsare needed. By measuring ΔV over a relatively long time period oneeliminates noise or transient factors from introducing errors. Althoughin FIG. 46 only two stages are shown, switching power supply controller1200 may be controlling a plurality of circuits which are powered byBatt. Ø. It will be understood that this method of measurement is onlyappropriate when all power outputs being driven by Batt. Ø are in lowcurrent mode. If one output were in a high-current mode and others inlow power (e.g., sleep) mode, the outputs in low power mode would simplybe ignored, a very small error compared with the energy utilized by thehigh power stage or stages.

In the above technique, the charge removed from the capacitor ismeasured over time. As described elsewhere, the system is informed aboutthe characteristics of the external components connected to it. Thesystem uses these values of the external components and utilizes them inthe coulombmetric calculation. This is important because since the exactcapacity of the battery is unknown, we measure how much goes in, wemeasure how much goes out, then correct for temperature and otherfactors which affect battery capacity.

Battery capacity varies from unit to unit, so there is no way to knowthe capacity in one cycle. Products that utilize coulombmetrics aretypically inaccurate until they go through a charge/discharge cycle. Inthe method of the present invention, in low current mode, we measurevery, very low current without any efficiency loss which is suffered inthe prior art by having to use large values of resistance for the senseresistor. We also avoid the use of a very high resolution A to Dconverter because we are resolving time in our implementation. Time isthe factor where we have the most precision of all. The value ofresistors R1 and RØ depend upon the supply design, not the coulombmetricneeds. They would typically be in the range of 20 to 100 milliohms.

In accordance with the present invention for coulombmetric measurementin conditions of high and medium power delivery, a measure is made ofthe power delivered to the load. This will be equal to the power removedfrom the battery, scaled by efficiency. This technique is also usefulfor measuring the energy put into the battery during charging; the sametechnique is used, the battery simply being the load. The advantage isthat what is measured is the net energy delivered to the battery by theexternal supply, allowing the designer to ignore power delivered duringthat time to the other loads. Thus the estimate of total power availablefrom the battery remains one of keeping a running total of net energystored in the battery, considering temperature and other factorspreviously discussed.

Consider now the high current usage case. Current must be measured on acycle by cycle basis. This is because power is delivered to the load insome portion of every 2 μsec cycle, with the duty cycle beingcontinuously recalculated and set to hold the voltage V₀ within thedesired controlled interval. Referring again to FIG. 46, one solutionwould be to measure the total current delivered by the battery Batt. Øby measuring the voltage across resistor RØ. However, to sample outtransients and noise would require frequent sampling throughout eachcycle, especially in systems with multiple outputs active. This wouldcreate a computing burden that would be equal to that of regulationitself. Instead, according to one technique of the present invention, adetermination is made of the power removed from the battery for eachcycle, using only one summing the power over many cycles.

A simplified implementation is described below using FIG. 46B in theexplanation. This scenario has a baseline current of 2 amps with aripple current of 200 milliamps, as depicted in FIG. 46A. That is, thetotal current swings from 1.9 to 2.1 amps. The technique of theinvention is to sample the current at its peak, which is known to occurjust before FET1 switches off and FET2 switches on. This is done bysampling the voltage across the output stage via sense lines S3 and S4using sense resistor R3. Although sense resistor R3 has a low value (asdiscussed in the low current mode description), current I is now highenough to cause an adequate voltage drop for A to D conversion.

The on time is known, the off time is known, therefore dt is known forthis stage. We know the output voltage (V_(o)) which is measured usingsense line S4 and the input voltage (V_(in)=SupplyA), thus we know thevoltage across the inductor L3 (V_(in)−V_(out)) and we know thecharacteristics of inductor L3. This tells us how much energy is beingdelivered to the load on a 2 microsecond basis. Typically, the systemreports coulombmetric information out ten times per second, a very fastupdate rate for coulombmetrics. By keeping a running total of powerdelivered to the load and knowing the efficiency and othercharacteristics of the battery, we have a very precise measure of thepower available from the battery at any moment. So for high currentscases, coulombmetric measurements are available from the method that weuse to perform regulation. No additional circuitry is required, only acalculations. In the prior art, one of the major jobs is to filter outnoise. The system of the present invention synchronously samples withthe noise source so there is no noise to filter since it is not seen atsampling time.

In the continuous mode, which is illustrated in FIG. 46D,(T_(Drive)+T_(Sync))=T_(Total)=2 μSec. As used herein, the phrase“continuous mode” means that current is flowing through the inductor forthe entire switching cycle. In a discontinuous mode, which isillustrated in FIG. 46C, the relationship is:(T_(Drive)+T_(Sync))<T_(Total).

Referring to FIG. 46D, an example is illustrated of the current Ithrough inductor L3 as a function of time over a 2 microsecond period,illustrative of continuous mode. In this example, a peak current of 200milliamps, indicated by I_(Pk), is achieved in one microsecond. Duringthe second microsecond of the period the current declines to zero.During the first half of the period, FET1 is conducting and FET2 isnon-conducting. During the second half of this period the conductionreverses. In this example, in the second half of this period the currentthrough inductor L3 has declined to zero, but has not reversed, as isthe case in some instances described herein. The period during whichFET1 is conducting is indicated in FIG. 46D as T_(Drive) and the periodduring which FET2 is conducting as indicated by T_(Sync). The total timefor the measurement period is indicated by T_(Total).

Now assume the system to be operating in the discontinuous mode asillustrated in FIG. 46C. The load requires an intermediate current, forexample 25 ma. The time that it takes for capacitor C3 to decay isimmaterial because a pulse is being supplied in every cycle. We want apeak current of 100 ma and an average current during the first phase,from time 0 to 1 μsecond, of 50 ma and a total output current of 25 mafor the 2 μsec period. Calculating 100 ma using R3 is difficult becausethe value of R3 is low, typically 0.05 ohms, and accordingly we havevery little resolution. However, in the circuit, we know that the valueof inductor L3 is 20 μh in this example. We know precisely what theinput voltage is (V_(in)=8V), what the output voltage is (V_(out)=4V),the time it was on (1 μsec) and that gives us the current that went intoinductor L3, the time that it was discharged and this gives us thecurrent that was flowing out of inductor L3 and we know the time it wasoff. Again, we can precisely calculate the load on a cycle by cyclebasis.

If the output voltage V_(out) were lower, it would take much less timeto charge inductor L3 than to discharge it. If (V_(out)) were, forexample, 2 volts, we would have 6 volts across inductor L3 and thecurrent would ramp up more rapidly than down. From the foregoing, itwill be appreciated that all the numbers required to calculate thecurrent are available without actually directly measuring it. We providebelow a general equation for calculating the current going into theload. Instead of measuring current as is done in the prior art, onlytime and voltage measurements are required. The system takes thesemeasurements at a rate of 500,000 times per second, however,coulombmetric data is only updated 10 times per second by taking thesenumbers continually and averaging them.

FIG. 46E illustrates an example in which the coil current of inductor L3is 100 milliamps at the beginning of the drive period and at the end ofthe drive period with the current flow in the direction of L3 to R3.During time T_(Drive), FET1 is conducting and FET2 is non-conducting andduring the T_(Sync) period the conduction of the FETs is reversed. Thetotal time period is indicated by T_(TOTAL). In this example, peakcurrent I_(Pk) is greater than ripple current I_(Ripple). As will beappreciated, I_(Ripple) is equal to 200 milliamps and I_(Pk) is 300milliamps. This, of course, results in (I_(Pk)−I_(Ripple))=100milliamps.

The average current per second flowing during the time periodsillustrated in the above examples maybe determined by the formula:${I_{\text{Avg}\text{/}\text{s}\quad\text{ec}} = {{\frac{V_{i\quad n} - V_{out}}{L} \cdot \frac{1}{2} \cdot \frac{T_{DRIVE}}{T_{TOTAL}}} + \quad{\frac{Ø - V_{out}}{L} \cdot \frac{1}{2} \cdot \frac{T_{SYNC}}{T_{TOTAL}}} + I_{Pk} - I_{Ripple}}}\quad$where T_(SYNC) is the time required for the inductor current to reachzero in the discontinuous mode or (T_(Total)−T_(Drive)) in continuousmode. Also, for the calculation above, it is assumed that the currentthrough inductor L3 is not allowed to go negative (i.e., flow fromV_(out) toward terminal S); and that the peak current, I_(Pk), throughinductor L3 is less than its saturation current.

The first term in the above equation represents the A portion of thecurrent contribution indicated on the figures, the second term of theequation represents the B portion of the current indicated in thefigures and the final term is indicated by C which is the continuingcurrent that flows in inductor L3. It will be appreciated from the abovethat the current being delivered can be calculated using informationavailable in the system rather than being measured as is the case in theprior art. This greatly simplifies the task, as well as reducing thecircuitry required.

Note now an optional variant of discontinuous mode. As power demand froma stage is reduced, the duty cycle, hence T_(DRIVE) time, is reduced.When this time becomes small the switching losses of the FETs can becomea significant loss of energy in the system. The same duty cycle may beachieved with less loss, hence better efficiency, by not turning on FET1for one or more cycle times, thus averaging T_(DRIVE) and T_(SYNC) overthis time which is now longer than (but a multiple of) 2 μsec. This modeis called “the cycle skipping mode.” Note, then, that the above formulastill applies, with T_(TOTAL) equal to the time interval betweenT_(DRIVE) pulses. It will be recognized, then, that continuous anddiscontinuous modes are simply a special case wherein T_(TOTAL)=2 μsec.

Section 1.4.2 Total Energy Calculation from Multiple Channel Data

Consider an array of switching power converters as shown in FIG. 47.This is an exemplary arrangement; there could be many otherarchitectures. As stated earlier, any output stage may be freelyconfigured as a buck converter (for example, stages SP1, SP2, SP3, SP4),or as a boost converter (for example, stages SP5, SP6). Stages SP5 orSP6, then, when connected to a battery, may simply connect the batteryto a supply bus SB #1 if the battery voltage is above what is needed forthe buck converters SP1-SP4. Alternately, switching power supplycontroller 1200 can configure converters SP5 and SP6 as boost converterswhen the voltage of the batteries are below what is required to meet theneeds of the buck converters, but above the minimum useable energy ofthe battery. Additionally, switching power supply controller 1200controls the charging of the batteries if charging is needed and anexternal supply is connected to bus SB1. Switching power supplycontroller 1200 selects between BAT5 and BAT6, depending upon the needsof the system and the energy available in each battery. For ease ofillustration, less than all of the control connections between switchingpower supply controller 1200 and converters SP1-SP6 are illustrated. Afull set of connections between switching power supply controller 1200and controlled converters are illustrated in other portions of thisspecification.

In this multiple-output system the energy state of BAT5 and BAT6 (and ofthe two collectively) is determined by employing the coulombmetricmeasurement techniques discussed herein, that is, determining the energydelivered by each of the converters SP1, SP2, SP3, and SP4, factoring inthe efficiency of the system, as well as certain battery parameters asdetailed elsewhere. Then the power profile of the whole system is thesum of these elements. This information is reported to switching powersupply controller 1200 for purposes of load balancing, load shedding,tuning an individual channel or system for optimal efficiency, oridentification of a problem or even failure.

Once this is understood, it will be appreciated that an array of systemssuch as that in FIG. 47 may be utilized and the information/controlmanaged by a higher level control system. One example is a telephonecentral office, where the system can detect a problem in an area (e.g.,cooling suddenly blocked to a specific area) or an individual card orsupply channel (e.g., a short or open in one of it's loads). As before,no additional hardware is needed, simply reuse of known data for anotherpurpose.

Section 1.4.3 Battery Life-time Estimate in an SPS

The above details the invention's determination of energy removed from abattery. To complete the system description, refer to FIG. 48. Thesystem includes a battery source indicated as Batt. 1.4.3; a chargerboost circuit CB1, which can function as a charger when external voltageis provided, that is the battery can be a load to the CB1 supply and theenergy provided to battery 1.4.3 measured per the above descriptions;and a buck converter CB2. As explained more fully elsewhere in thespecification, circuit CB1 can operate as a switch when the batteryvoltage is above the voltage required by the buck converter CB2, or as aboost supply when the battery voltage is below the voltage required forthe buck supply CB2. Both of these power supplies are monitoring theenergy that is flowing through them and provide that information to theprocessing element 1.5 which can be part of switching power supplycontroller 1200. Processing element 1.5 also receives temperature datafrom temperature sensor element T.

Temperature sensor T may be a thermocouple, a thermal variable resistor,or it could be a Kelvin temperature sensor. In one implementation of theinvention, a Kelvin temperature sensor is used. The integrated circuitalso has pins to support an external Kelvin temperature sensor,depending upon the proximity of the integrated circuit to the battery.Data from one or the other of the sensors, or both, can be used. Duringcharge cycles, the total amount of energy that is provided to thebattery is monitored, scaled with the battery temperature using dataprovided by the battery manufacturer, and the total charge that thebattery absorbs is computed. Batteries absorb more charge when they arehot than they do when they are cold. Also, they have more energyavailable when they are discharged hot than when they are cold. Theworst possible combination is to charge it cold and use it cold. Byusing capacity degrade data to first compute how much energy is actuallydelivered to the battery, then monitoring the temperature and the rateat which the charge is being removed from the battery by the buckconverter CB2, which reports its power information to processing element1.5, the time remaining until battery exhaustion is calculated.

FIG. 48A illustrates a typical battery degrade curve which plotscapacity (C) versus temperature (T). This information is published bybattery manufacturers. The battery degrade curve information may beprogrammed into processor 1.5.

In these applications the battery is known to the system. It is eitherpart of the system, or the power supply system is built into the batterymodule. The battery degrade information about the battery capacity canbe stored in nonvolatile memory indicated by NVM1 in the figure. It'salso true that for families of battery chemistries there are similarcharacteristics. So although the given capacity for some manufacturersare better or worse, the decay rates, for example for nickel metalhydride batteries, are similar even if one does not know the data forthat specific brand of battery. Clearly, the more stable the temperatureis, the less important the degradation data is.

Section 1.4.4 SPS Current·Voltage (Power) Regulation Based onCoulombmetric Data

In the previously described embodiment the coulombmetric methodology maybe used to accumulate and report energy consumption or input.

An alternative implementation is for the coulombmetric data to be usedas an input for regulation. With cycle by cycle energy as the regulatedparameter in a control loop constant energy could be delivered to a loador consumed from a source. This regulation could be accomplished byadjustment on a cycle by cycle basis of the duty cycle of the FETs basedon maintaining a constant coulombmetric (energy) value. One applicationof this technique would be to regulate the input energy to a radiofrequency power amplifier for the purpose of power level control.Another would be to regulate the power delivered by a photovoltaic cell.

In the previously described embodiment the coulombmetric methodologymade use of certain voltage measurement obtained for use by theregulation control program. That is, the coulombmetric algorithm programreuses voltage measurements (for example, the voltage across senseresistor R1 in FIG. 46) taken by the voltage regulation program andstored into a predetermined memory location.

An alternative implementation is for the coulombmetric program loop totake this data instead, and the regulation algorithm program can thenmake use of the stored data. An advantage of the current invention isthe ability to take data only once, then make the data available forother purposes. By using the program control, the data of interest istaken during each cycle frame, and that the data is saved for reuse byother program(s).

The voltage control program (if that is the preferred method of controlfor a given output channel), then, would make use of said data measuredwithin the coulombmetric loop to control voltage as previouslydescribed. That is, adjustment on a cycle by cycle basis of the dutycycle of the FETs.

Section 1.4.5 SPS Constant Energy Output Regulation Mode

Previous descriptions of the application of the invention have focusedon voltage, controlled to remain within a certain target range. Havingthe capability to collect and utilize coulombmetric data permits its useto instead control power directly. That is, the combination of voltageand current, not simply one or the other. This is beneficial forapplications where power must be kept within a target range, such as theRF output of a cell phone, or where control of the power level can leadto optimized efficiency, such as with a solar panel array. Recall thatthe invention develops this information with fine granularity andaccuracy on a cycle by cycle basis. This information may then be used tomanage the control loop.

A highly simplified example of a typical cell phone is shown in FIG. 49.The desire for best performance while keeping output below anagency-imposed maximum suggests the value of controlling to a specificpower level, say, 1 watt. Power supply 1.4A may be a buck or boosttopology, and typically the power amplifier 1.4 would have an efficiencyof 50%. The strategy then is to control the output of Supply 1.4A toprovide a constant 2 watts into power amplifier 1.4, compensating forchanges in voltage or current, depending upon temperature, and unit tounit variations. This is accomplished by using switching power supplycontroller 1200 to regulate power-supply 1.4A.

For another example, consider FIG. 49A. The system includes an array ofphotovoltaic cells 1.4.3, which are essentially large silicon diodes.These diodes produce current at a voltage that is dependant upontemperature, influenced by approximately −2.1 mV per degree C. Therewould typically be one hundred such diodes in series, making the effectof temperature significant. The system may be arbitrarily operated atany of a range of voltage/current combinations, but only one combinationproduces the maximum power and that specific combination changes withtemperature. The maximum power is that combination that maximizes thearea under the V-I characteristic curve. Importantly, the need is tomatch the power delivered from the (solar array) source at theconditions that optimize the optional battery's charge acceptance rateand/or tuning for maximum power delivery to the grid.

One technique for determining the optimal set point is to make smallchanges in the output of buck converter 1.4.3.1 via switching powersupply controller 1200, examining the resulting power usingcoulombmetric methods already discussed, then comparing the result tothe previous power level and selecting that which is highest. Theprocess is repeated, maintaining the array at its peak. As before, nonew hardware is required for this procedure.

Section 1.4.6 Charge Decay Time Energy Extrapolation

Referring to FIG. 41, regulation control module (REG) 1204 includes acoulombmetric and temperature module 600.1. Because regulation controlmodule (REG) 1204 receives voltage and current feedback information fromthe switching power converter, it can count the number of coulombs ofcharge transferred from supply batteries to the loads powered by theswitching power converters. This provides a more accurate estimate ofremaining battery life than by merely using the current battery voltageto estimate remaining battery life. As described further with respect tocentral processing module (SYS) 1205, the circuit parameters such asinductance values and capacitance values for each of the switching powerconverters may be stored in memory within central processing module(SYS) 1205. By processing the feedback information in light of thesecircuit parameters, coulombmetric and temperature module 600.1 maydetermine the amount of coulombs supplied by a battery. For example, ifthe feedback voltage from a switching power converter measures thevoltage across a load capacitor, the amount of charge drawn through thecapacitor is:ΔQ=C*ΔVWhere C is the capacitance of the load capacitor, ΔV is the change involtage as determined from the voltage feedback samples, and ΔQ is theamount of charge transferred to the load. It will be appreciated thatthe above example is illustrative only and that coulombmetric andtemperature module 600.1 may determine the amount of charge transferredfrom a supply battery in other ways.

Coulombmetric and temperature module 600.1 may store the resultingcharge measurements for central processing module (SYS) 1205 to accessthrough bus interface 525.1.

Section 1.5 Cyclic Switching Frequency Modulation

As described herein in section 1.1.4 and referring to FIG. 37 and FIG.37B, a CAM 2486.4 illustrated may generate signals indicating pulserising and falling edges for a number of independent pulse channels asspecified by data from Regulation control module (REG). Each pulsechannel includes a number of pulses for controlling a correspondingexternal pulse-width-modulated (PWM) switching power converter. Duringstandard operation, CAM 2486.4 receives a read command in the form of acount from a counter, for example, a Grey counter 2484.4, that commandsCAM 2486.4 to check each possible data storage location to see ifregulation control module (REG) 1204 has written a data word in a givendata storage location matching the current count.

In one embodiment, CAM 2486.4 has storage for 64 data words. These 64data words correspond to 8 pulse channels, wherein each pulse channeldefines 4 pulses. Seven of the pulse channels are used for externalPWM-switching power conversion and the eighth pulse channel is used togenerate auxiliary pulses. For the seven pulse channels, 56 data wordsneed be specified. The remaining eight data words correspond to the 4auxiliary pulses edges of an auxiliary signal AUX 2446.4 used forexample to synchronize external circuitry.

As Grey counter 2484.4 counts through a complete cycle, it causes CAM2486.4 to check its stored data words for any matches. Each cycle ofGrey counter 2484.4 defines a single DPC frame as shown in FIG. 37. Inturn, it can be seen that the clocking of Grey counter 2484.4 controlsthe minimum distance between possible pulse rising and falling edges.

For example, suppose Grey counter 2484.4 is a 10-bit counter andreceives a 268.4 KHz clock signal. If Grey counter 2484.4 is configuredto count at both the rising and falling edges of the clock signal, Greycounter 2484.4 will then count at a 536.9 MHz rate. The resulting DPCframe rate, which equals the count rate divided by the maximum count,will be 524.3 KHz (536.9 MHz/1024), wherein each DPC frame is dividedinto 1024 possible pulse rising and falling edge locations separated bya duration of approximately 2 nanoseconds. The division of each DPCframe into these possible rising and falling edge locations determinesthe minimum pulse-width-modulation (PWM) resolution for any PWMswitching power converter being controlled by a given pulse channel.Thus, the count rate for Grey counter 2484.4 determines the minimum PWMresolution as given by the inverse of the count rate.

As can be seen in FIG. 37B, the count rate for Grey counter 2484.4 willultimately depend on DPLL clock signal PLLCK 2460.4 from DPLL 2480.4.Because DPLL clock signal PLLCK 2460.4 is used for multiple purposesthroughout switching power supply controller 1200, such as a relativelyfast clock necessary for analog to digital converter 1206, PLLCK 2460.4having a frequency of 536 MHz may be divided in spreader divider 2482.4to clock Grey counter 2484.4 at a lower clock frequency of, e.g., 268MHz.

Regardless of the specific frequency Grey counter 2484.4 is clocked at,the count rate and the number of bits used in Grey counter 2484.4determines the DPC frame rate and the minimum PWM resolution. The risingand falling pulse edges for a plurality of external PWM-switching powerconverters are determined with respect to this minimum PWM resolution.The FET switches in each PWM-switching power converter will switch onand off at the DPC frame rate. Accordingly, each PWM-switching powerconverter will produce RF noise at the DPC frame rate frequency.

To assist in the reduction of this RF switching noise, spreader divider2482.4 includes a divider 200.1 and a spectral spreader 210.1 as shownin FIG. 50. Divider 200.1 receives DPLL clock signal PLLCK 2460.4 fromDPLL 2480.4 and provides a divided clock signal 215.1 to spectralspreader 210.1. Spectral spreader 210.1 dithers the frame rate so as tospread the RF switching noise in a spread spectrum fashion. By skippingor “swallowing” various clock cycles received from divider 200.1,spectral spreader 210.1 may achieve this dithering. In turn, spectralspreader 210.1 will skip clock cycles in divided clock signal 215.1 toproduce a desired amount of dithering to the DPC frame rate by providingan adjusted clock signal 220.1 to Grey counter 2484.4.

An exemplary embodiment for spectral spreader 210.1 is shown in FIG.50A. A divider 300.1 and a counter 310.1 receive divided clock signal215.1. Divider 300.1 provides a secondary divided clock signal 320.1 toan up/down counter 330.1, which in turn provides an up/down count 335.1to control a multiplexer 340.1. Multiplexer 340.1 selects bits from acount 345.1 produced by counter 310.1 As controlled by up/down count335.1, multiplexer 340.1 chooses either the most significant bit orsuccessively less significant bits from count 345.1 to provide aselected bit 355.1 to the D input of a D-type flip-flop 350.1 that isclocked by divided clock signal 215.1. A NAND gate 360.1 receives bothselected bit 355.1 and a {overscore (Q)} output 365.1 of flip-flop350.1. Finally, an AND gate 370.1 receives an output 375.1 from NANDgate 360.1 and divided clock signal 215.1 and outputs adjusted clocksignal 220.1.

Accordingly, if NAND output 375.1 is true, the cycles of adjusted clocksignal 220.1 will correspond to the cycles of divided clock signal215.1, that is, no skipping occurs. However, if NAND output is falseduring a given cycle in divided clock signal 215.1, this cycle will beskipped in divided clock signal 215.1. It will be appreciated that thenumber of skipped cycles, and hence the amount of spectral spreadingdepends upon the sizes of counter 310.1 and up/down counter 330.1 andthe division provided by divider 300.1. For example, suppose dividedclock signal is 268 MHz, divider 300.1 divides by 1024, up/down counter330.1 is a 3-bit counter, and counter 310.1 is a 15 bit counter(corresponding to the 2 nanosecond PWM resolution discussed previously).Secondary divided clock signal 320.1 thus cycles according to the DPCframe rate. Initially, up/down count will be at zero such thatmultiplexer 340.1 selects the most significant bit in count 345.1. Whenup/down count 335.1 increments, multiplexer 340.1 will then select thenext-most significant bit and so on. It follows that the following pulseskipping schedule will be followed:

1 pulse in 32 frames skipped (in a duration of 32 frames)

1 pulse in 16 frames skipped (during the next 16 frames)

1 pulse in 8 frames skipped (during the next 8 frames)

1 pulse in 4 frames skipped (during the next 4 frames)

1 pulse in 2 frames skipped (during the next 2 frames)

2 pulses in 1 frame skipped (during the next frame)

4 pulses in 1 frame skipped (during the next frame)

2 pulses in 1 frame skipped (during the next frame)

1 pulse in 2 frames skipped (during the next 2 frames)

1 pulse in 4 frames skipped (during the next 4 frames)

1 pulse in 8 frames skipped (during the next 8 frames)

1 pulse in 16 frames skipped (during the next 16 frames)

1 pulse in 32 frames skipped (in a duration of 32 frames)

whereupon the entire pulse-skipping schedule would be repeated. As aresult, the DPC frame rate will vary in a nonlinear fashion. It will beappreciated that numerous other pulse skipping schedules could beimplemented using alternative embodiments of spectral spreader 210.1.For example, the DPC frame rate could be varied in a linear fashion.

Section 1.6 PS/PM/Fault Management Integration—See Disclosure in DataSheet

Section 1.6.1 Load Shedding in an SPS

Central processing module (SYS) 1205 may be initialized by a hostprocessor with the operating parameters and topology for a plurality ofswitching power converters under the control of switching power supplycontroller 1200. These operating parameters may include operatingthresholds such that central processing module (SYS) 1205 will cause theoperation of a given switching power converter to cease if thecorresponding thresholds are not satisfied. These operating thresholdsmay comprise a minimum supply voltage required for a given switchingpower converter or the minimum amount of charge remaining in a batteryor set of batteries used to power the switching power converter.

Microprocessor core 400.1 (FIG. 56) in central processing module (SYS)1205 monitors these operating thresholds and responds accordingly. Forexample, microprocessor core 400.1 receives an interrupt periodically toupdate coulomb-metric measurements and battery temperature readings withrespect to the power supplies powering the various switching powerconverters under control of switching power supply controller 1200. Inconjunction with these coulomb-metric updates, microprocessor core 400.1may check that the amount of charge remaining in the battery or set ofbatteries is sufficient to sustain the respective supplies until thenext checking interval. In conjunction with the coulomb-metric updates,microprocessor core 400.1 may also check that the associated supplyvoltages satisfy the voltage minimums for the various switching powerconverters.

Should the amount of charge remaining/or the supply voltage not satisfythe specified minimum for a given switching power converter,microprocessor core 400.1 uses a configured deterministic algorithm andcommands regulation control module (REG) 1204 to cease operation of theappropriate switching power converter. At the same time, microprocessorcore 400.1 notifies the host processor (not illustrated) that theparticular switching power converter is being brought down. In thisfashion, a “load shedding” is accomplished with respect to the loadspowered by the switching power converters being brought down. It will beappreciated that this manner of load shedding is efficient as comparedto a conventional method of using the host processor to monitor variousswitching power converters and directly commanding them to turn offshould an operating threshold be exceeded because a host processor willtypically demand far more power than microprocessor 400.1 would inresponding to interrupts to check the operating thresholds.

Section 1.6.2 Power Cycling in an SPS

Upon expiration of Host watchdog timer 1.1 shown in FIG. 53, a resetsignal/command is issued to a host microprocessor (not illustrated).However, in certain malfunctions such as latch-up, a host microprocessorwill not respond to a reset status notification or physical reset signalassertion. Instead, a latched-up host microprocessor will continue todraw current until its power source is exhausted or shut off. Assumingthat its transistors have not been damaged by the latch-up, the hostmicroprocessor may then be reset.

Central processing module (SYS) 1205 has the ability to provide anintelligent response to such malfunctions. Once a host watchdog expiresthe host is notified by either a physical reset assertion or a resetstatus notification (via an Interrupt and command response). The Centralprocessing module then restarts the host watchdog and if the host doesnot attempt to enable (kick) the watchdog within a configurable numberof watchdog expirations the power supplies associated with the host(configurable) is cycled off for a configurable time duration andrestarted. For example, a certain voltage level may be needed for thehost CPU's memory, another for its input/output circuitry, and otherlevels for the CPU itself, where each voltage level is provided by aswitching power converter under the control of switching power supplycontroller 1200. Stored in the Non Volatile Memory are the correct powersequencing for the host CPU's switching power converters for both apower up and a power down sequence. After power down has been completed,central processing module (SYS) 1205 may then power up the host CPU byappropriately sequencing the involved power converters.

Upon expiration of 8051 watchdog timer 5.1 shown in FIG. 53, a reset maybe issued to the internal microprocessor (not illustrated) causing awarm boot of the internal microprocessor. This reset may not cause theregulation of power to be interrupted rather it merely restarts theinternal microprocessor. Additionally there may be a status bitindicating a watchdog event occurred for the internal microprocessor toread upon rebooting to allow it to ascertain the reason for booting.

Section 1.6.3 Reset Conditioning in an SPS

As described with respect to watchdog controller 1213 of FIG. 53,switching power supply controller 1200 may reset a host CPU (notillustrated) by asserting a reset signal if Host watchdog timer 1.1expires. Because of the intelligent control provided by centralprocessing module (SYS) 1205, the reset may be asserted until certainconditions are satisfied. For example, regulation control module (REG)1204 may signal to central processing module (SYS) 1205 that certainswitching power converters are producing voltage levels outside of adesired operating range. For example, the output voltage of a switchingpower supply providing power to the memory of the host CPU may be out ofrange causing the host to not provide the appropriate resetcommand/signal. In such a case, central processing module (SYS) 1205could continue to assert the reset signal until all power suppliesaffecting operation of the host CPU are providing voltages withindesired operating ranges.

Section 1.6.4 and x.3, Watch Dog Structure and Watch Dog Timer in an SPS

Because switching power supply controller 1200 may supply power to ahost CPU-containing device such as a personal digital assistant (PDA),watchdog controller block 1213 (FIG. 12) may provide a watchdog featurefor the host CPU. A common problem for operating systems running on CPUsis a “lock-up” condition resulting from conflicting program commands,invalid memory access requests, and related matters. Recovery from suchproblems may take an extended, unknown amount of time or may not occur.Accordingly, watchdog timers comprising a digital counter that countsdown to zero from a predetermined starting number are often implementedin CPUs to avoid lock-ups. During normal operation, the CPU willperiodically command the watchdog timer to reset the counter to avoidthe timer's expiration. However, in a lock-up, the CPU will not commandthe watchdog timer to reset the counter whereby the timer expires. Inresponse, a reset may then be initiated to remove the lock-up.

Because switching power supply controller 1200 includes centralprocessing module (SYS) 1205, the watchdog controller block 1213includes two types of watchdog timers as seen in FIG. 53: a hostwatchdog timer 1.1 for an external host CPU and a 8051 watchdog timer5.1 for the internal mocroprocessor.

If host watchdog timer 1.1 expires, watchdog controller 1213 notifiescentral processing module (SYS) 1205 via an interrupt. In response tothis interrupt, central processing module (SYS) 1205 either asserts areset signal to the host or asserts an interrupt line to the host as isdictated by a configuration parameter.

If 8051 watchdog timer 1.1 expires, watchdog controller 1213 will assertthe reset line central processing module (SYS) 1205. This triggers awarm boot of the internal microprocessor.

Watchdog controller 1213 may have two modes of operation: normal modeand power save mode. In the normal mode, both host watchdog 1.1 timerand 8051 watchdog timer 5.1 are operative. In the power save mode, thehost microprocessor is required to disable the host watchdog prior toissuing a power save command unless there is not a communications linkbetween the host and the SPS upon which the Watchdog controller 1213will rely on a configuration parameter that specifies automatic hostwatchdog enabling/disabling upon application of or removal of powerassociated with the host.

Each watchdog timer 1.1 and 5.1 may associate with its own serviceregister (not illustrated) used to reset the timers. To reset itsparticular watchdog, central processing module (SYS) 1205 writes apredetermined code word and the inverse of that code word to arespective service register. An errant write to a service register doesnot reset the associated watchdog timer and generates an interrupt tocentral processing module (SYS) 1205. In addition, the host watchdogtimer 1.1 may have its counter reset based upon a toggled binary signalreceived from the host microprocessor. This signal is processed bycentral processing module (SYS) 1205 which in turn resets the watchdog.However, resets using the kick watchdog command are more secure and lesssusceptible to runaway conditions in the host microprocessor.

To provide greater control flexibility to the host processor, Hostwatchdog timer 1.1 and 8051 watchdog timer 5.1 may be initializedaccording to data stored in associated configuration registers (notillustrated). By writing to these registers, the duration of theindividual watchdog counter timeout periods can be configured. Forexample, this configuration could be 15 bits for the host watchdog toaccommodate a range of 1 ms to ˜32 seconds while 6 bits could be usedfor the 8051 watchdog to provide a range of 100 ms to 3.2 seconds. If agiven watchdog timer is not enabled, the watchdog timer will not bestarted until its configuration register is re-initialized. To preventerrant access of the 8051 watchdog configuration register, thisregisters may be written to only once after a reset operation (until thesubsequent reset operation, whereupon the configuration register may bere-initialized). The host watchdog register may not have thisrestriction.

As discussed above, the host watchdog may issue an interrupt to centralprocessing module (SYS) 1205 should host watchdog timer 1.1 expire. Awatchdog interrupt status register 15.1 may also store a bit to indicatethat either watchdog's service register has been serviced with an errantcodeword, potentially signaling an invalid memory access. A watchdoginterrupt mask register 15.1 (shown combined with the watchdog interruptstatus register in FIG. 53 for illustration clarity) may store bitsindicating whether the host watchdog timer has been masked. In thisfashion, central processing module (SYS) 1205 may prevent watchdogcontroller 1213 from generating interrupts during system criticalperiods. In addition, watchdog interrupt mask register 15.1 may store abit indicating whether interrupts resulting from an errant codewordwrite to the service register for Host watchdog 1.1 should be masked.

Section 1.6.5 Programmable Reset and Watchdog Functions

As described with respect to watchdog controller 1213 of FIG. 53switching power supply controller 1200 may reset a host microprocessor(not illustrated) if host watchdog timer 1.1 expires by asserting areset signal. Because of the intelligent control provided by centralprocessing module (SYS) 1205, the reset may be asserted until certainconditions are satisfied. For example, regulation control module (REG)1204 may signal to central processing module (SYS) 1205 that certainswitching power converters are producing voltage levels outside of adesired operating range. For example, the output voltage of a switchingpower supply providing power to the memory of the host microprocessormay be out of range. In such a case, central processing module (SYS)1205 could continue to assert the reset signal until all power suppliesaffecting operation of the host microprocessor are providing voltageswithin desired operating ranges.

Once all conditions have been satisfied, central processing module (SYS)1205 allows the reset to be de-asserted. However, even though all powersupplies are operating correctly, a host device may still not havestabilized properly to justify release of the reset command.Accordingly, the duration of the reset command after all conditions havebeen satisfied may be programmable. The duration may be stored bycentral processing module (SYS) 1205 as programmed by a user.

Section 1.6.6 Resistive Digitizer in Combination with an SPS

Quad-Slope Analog to Digital Converter 1211 a in one embodiment of thepresent invention measures the point of contact between two sheets ofresistive material. Touch screens suitable for implementing the presentinvention are commercially available from manufacturers such as 3M. FIG.18B is a highly simplified illustration of a four-contact touch-screen18.8 having a first sheet 18.81 and a second sheet 18.82. These sheetsare shown offset for ease of illustration, but are of course aligned fornormal use. Each sheet has a uniform sheet resistance such that thephysical point of contact along the surface of each sheet may berepresented by the proportion of end to end resistance at that point.Using sheet 18.82 and applying an electrical potential to terminals TOP18.83 and BOT 18.84 permits, using the other sheet as a contact sheet, adetermination of the point of contact in the Y direction. Applying anelectrical potential to terminals LFT 18.85 and RHT 18.86 using sheet18.81 as a sensing sheet, permits, using sheet 18.82 as a contact sheet,a determination of the point of contact in the X direction. Thus in eachcase a voltage is applied across one of the two sheets (e.g., 18.81 or18.82, the “sensing sheet”) while it is electrically isolated from QSADC1211 a, and simultaneously an ADC conversion is made of the voltagepresent at the point of contact on the other sheet (i.e., 18.82 or18.81, the “contact sheet”).

A voltage is applied to terminal LFT 18.85 and terminal RHT 18.86 of theX coordinate sheet 18.81, then the proportion of the applied voltage canbe read at either the TOP terminal 18.83 or BOT terminal 18.84 of the Ycoordinate sheet 18.82. The magnitude of the voltage read isrepresentative of the physical horizontal position of the point ofcontact between the two sheets. Similarly, if a voltage is applied toterminal TOP 18.83 and BOT 18.84 of the Y coordinate sheet 18.82, thenthe proportion of the applied voltage read at either terminal LFT 18.85or terminal RHT 18.86 of the X coordinate sheet 18.81 is representativeof the physical vertical position of the point of contact between thetwo sheets.

In another embodiment, one resistive sheet is used for both X and Ysensing while the other sheet is used to transfer the proportionalvoltage to the QSADC. A highly simplified illustration of a five-contacttouch screen 18.9 is shown in FIG. 18C. A highly simplified illustrationof a five-contact touch screen 18.9 is shown in FIG. 18C, and includessensing sheet 18.93 and contact sheet 18.92. Touching the front sheet18.92 causes 18.92 and 18.93 to contact each other. As in theillustration above, the sheets are shown off-set for ease ofillustration. In this embodiment where five connections are used, avoltage is imposed between the terminals LFT 18.94 and RHT 18.95 ofsheet 18.91, then the proportion of the voltage read at terminal 18.96of contact sheet 18.92 is proportional to the physical horizontalposition of the point of contact between the two sheets. Similarly, avoltage is imposed between the TOP terminal 18.97 and BOT terminal 18.98of sheet 18.93, then the proportion of the voltage read at terminal18.96 of contact sheet 18.92 indicates the physical vertical position ofthe point of contact between the two sheets.

Quad-Slope Analog to Digital Converter (QSADC) module 1211 a (FIG. 17)includes a custom low-power mixed-signal circuit that has both analogand digital input and output signals, including analog and digital powerand ground reference voltages. QSADC module 1211 a measures anddigitizes, to 8 bits of resolution, the voltages at two separateexternal ports relative to a voltage reference VREFH. In one embodiment,a conversion rate for QSADC module 1211 a is approximately 300 sps(samples per second).

FIG. 16 is a timing diagram illustrating the quad-slope (i.e., dualconversion) analog to digital conversion (ADC) operations carried out inthe QSADC module 1211 a. Time intervals 1601 and 1602 (i.e.,“measurement periods” PMA and PMB) are equivalent fixed time durationscorresponding to 256 counts of a reference timer or counter. During timeperiods 1601 and 1602, the positive slopes of signal trace segments 1603and 1604 reach voltages V_(Y) and V_(X), plus the small initial voltage,which represent the integration of the signals being measured on thecontact sheets by the QSADC module 1211 a over time periods 1601 and1602.

During time intervals 1605 and 1606 (“conversion periods” PCA and PCB),signal trace segments 1607 and 1608 have identical negative slopes,tracing the voltages as they decline from voltages V_(Y) and V_(X) tozero. During these conversion periods (PCA and PCB), the numericalcounts represent direct analog to digital conversions of the measuredvoltages from contact sheets 18.82 and 18.81, respectively. At thecompletions of the conversion periods, end-of-conversion (EOC) pulsesare generated (e.g., at times t₁ and t₂) to signal the end of eachanalog to digital conversion and to reset and zero the offsets of theanalog circuitry in QSADC module 1211 a to prepare for subsequentconversions.

FIG. 17 is a block diagram showing the interface signals of QSADC module1211 a in one implementation. As shown in FIG. 17, QSADC module 1211 aincludes: (a) Analog I/O interface (AIO) 1701, Digital Interface (DI)11702, and the Power Supply Interface 1703. Analog I/O interface (AIO)comprises analog bi-directional measurement terminals 1701 a and 1701 b(i.e., terminals TOP and BOT), analog bi-directional measurementterminals 1701 c and 1701 d (i.e., terminals RHT and LFT), analogreference capacitor ports 1704 a and 1704 b (PR1 and PR2, respectively),analog bi-directional measurement port 1701 f for the MID input, andReference Voltage signal 1701 e (i.e., reference voltage VREFH).

Analog I/O interface 1701 operates in two modes. In a first mode(“primary mode”) a port (e.g., Port X or Port Y) measures voltage by thetechnique described herein. Port X is comprised of switches A0, A1, A2and A3 (FIG. 18A). Port Y is comprised of B0, B1, B2 and B3 (FIG. 18A).In a second mode (“sleep mode”) ports X and Y are configured to detectelectrical continuity between them. A determination of continuitybetween Port X and Port Y indicates contact with the touch screen, whichthen causes the analog I/O interface 1701 to enter primary mode.

Digital interface 1702 comprises 8-bit output buses 1702 a (DOUTX[7:0])and 1702 b (DOUTY[7:0]) for displaying digital results of theconversions at ports X and Y, respectively, completion or “DONE” signal1702 c, Continuity Detection or “CONT” signal 1702 d, Start Conversionor Reset “START/RSTN” signal 1702 e, 128 KHz 50% duty cycle referenceclock signal (“CLK”) 1702 f, and Mode Select signal SEL 1702 g.START/RSTN is the start control pin used to initiate the analog todigital conversion processes. When the START/RSTN pin goes HIGH theprocessing starts and subsequent conversions will not be initiated untilthe START/RSTN pin toggles LOW then HIGH again. When START/RSTN is heldLOW, all necessary circuits in the block are held in a low power resetstate. Signal SEL 1702 g is received from touch screen interface 1211,and is used to control the ADC conversion mode. When the SEL 1702 g portis LOW the MID port is not used for ADC measurements. That is, afour-terminal arrangement for measurement of X-Y position is employed,as shown in FIG. 18C. When the SEL 1702 g port is held HIGH the MID portis used for measuring the input voltage at the X and Y ports. That is, afive-terminal arrangement for measurement of the X-Y position isemployed, as shown in FIG. 18C.

The DONE signal at terminal 1702 c is asserted when the digitalconversions at both ports X and Y are completed, to signal that resultscan be read from output buses 1702 a and 1702 b (i.e., DOUTX and DOUTYbuses). The CONT signal indicates detection of continuity between portsX and Y.

Power supply interface 1703 comprises analog power and ground referencesignals 1703 a and 1703 b (i.e., analog power and ground referencesignals AVD and AVS, respectively), and digital power and groundreference signals 1703 c and 1703 d (i.e., digital power and groundreference signals VDD and VSS, respectively).

FIG. 18 and FIG. 18A show one implementation of QSADC module 1211 a, inaccordance with the present invention. FIG. 18 is a top-level blockdiagram of QSADC module 1211 a, including analog (ANLG) block 1801,digital controller (CNTRL) block 1802, pre-settable up/down counterblock 1803, 8-input NAND gate 1806, and 8-bit Registers 1804 and 1805(i.e., registers REGX and REGY, respectively).

Table 2.6 provides a brief descriptive summary of the signals shown inFIG. 18.

TABLE 1.6.6a Port Name Type Description Source Destination TOP AnalogMax Analog Measurement Port AIO BOT Analog Min Analog Measurement PortAIO RHT Analog Max Analog Measurement Port AIO LFT Analog Min AnalogMeasurement Port AIO MID Analog Analog Measurement Port for the AIO MIDInput PR1 Analog External Reference Capacitor AIO First Port PR2 AnalogExternal Reference Capacitor AIO Second Port START/RSTN Input StartConversion Port Active TSI HIGH SEL Input Mode Select Conversation PortTSI QSADC_CLK Input Clock CLKGEN DOUTX[7:0] Output Digitized Bus ForMeasured TSI Values A DOUTY[7:0] Output Digitized Bus For Measure ValuesB TSI DONE Output Done Port Active HIGH TSI CONT Output ContinuityDetection Port Active TSI HIGH VREFH Input High Analog Voltage ReferenceIVS VREFL Input Low Analog Voltage Reference IVS AVD Power Analog PowerIVS AVS Power Analog Ground IVS VDD Power Digital Power IVS VSS PowerDigital Ground IVS

FIG. 18A shows one implementation of ANLG 1801 of FIG. 18, whichincludes operational amplifier 1851 and analog comparator 1852, inaddition to MOS transmission gates 1853 a to 1853 p, inverter 1854, ANDgate 1856, Digital Control block (DCNTL) 1855, and Level Shifter 1856.The common mode ranges of the operational amplifier 1851 and analogcomparator 1852 are rail to rail (i.e., 0 to 3.3V).

DCNTL block 1855 generates the control signals necessary to control theMOS transmission gates 1853 a to 1853 n. For convenience, the MOStransmission gates 1853 a to 1853 p are referred to as “switches” andeach switch is designated by its control input signal (i.e., A0-A3,B0-B3, MEAS, SLP, EOC, SEL, CONV, and CONV2).

Initially, QSADC module 1211 a is in sleep mode, where electricalcontinuity is checked. A four-terminal arrangement is illustrated byFIG. 51O. Switches 1853 b, 1853 c, 1853 h, 1853 i, 1853 n, 18531, and1853 f (i.e., signals A0, A2, B1, B2, MEAS and SLP) are closed, whileall other switches are open. When there is no electrical continuitybetween the two sheets, the output voltages of operational amplifier1851, analog comparator 1852, and the DOUT signal are all low. Afive-terminal implementation is illustrated in FIG. 51P. Switches 1853p, 1853 i, 1853 n, 18531 and 1853 f (i.e., signals B1, B2, SEL, MEAS andSLP) are closed while all others are open. When there is no electricalcontinuity between the two sheets, the output voltages of operationalamplifier 1851, analog comparator 1852, and the DOUT signal are all low.

When electrical continuity is established between ports Y and X, theoutput terminal of switch 1853 n (i.e., the non-inverting input terminalof operational amplifier 1851) is pulled to a high voltage, so that theoutput terminal of analog comparator 1852 is at a high voltage. Theoutput terminal of analog comparator 1852 is the “DOUT” terminal ofanalog block 1801 (FIG. 18), which is connected to the “DIN” terminal ofdigital control block 1802. In response to the high voltage at theoutput terminal of analog block 1801, the CONT terminal (i.e., terminal1702 d at FIG. 17) is driven to a high voltage, resulting in QSADCmodule 1211 a exiting the sleep mode, and entering the primary mode.

In primary mode (entered when the CONT signal is at a high voltage),when an active START/RSTN signal at terminal 1702 e is received fromtouch screen interface 1211, digital control block 1802 begins ameasurement cycle of the voltage at Port Y by presetting up/down counter1803 to hexadecimal FF (i.e., FFh), thereby resetting the CONT signal,and beginning the measurement and conversion process. Thereafter, eachrising edge of the CLK signal at terminal 1702 f decrements up/downcounter 1803 until the count reaches zero (for a total of 256 counts),which is decoded by NAND gate 1806 to activate the ZERO signal receivedinto digital control block 1802.

The operation of analog circuit 1801 is described below in connectionwith FIG. 51F through FIG. 51O. For convenience of illustration, a darkline is shown on the figures to denote the conduction path of thetransmission gates.

The measurement initial state is shown in FIG. 51F. In the initialcondition of the QSADC 1211 a prior to a measurement cycle, CNTRLcircuit 1802 drives EOC HIGH and MEAS LOW, thus asserting CONV frominverter 1854. Switch 1853 e connects the non-inverting input toamplifier 1851 to VREFL. Amplifier 1851 is connected to thenon-inverting input of amplifier 1852, which has its non-inverting inputpermanently connected to VREFL. The output of amplifier 1852 isconnected to the inverting input of amplifier 1851, completing anegative feed back loop spanning both amplifiers. Amplifier 1851 bringsits inverting input to the same potential as VREFL (the current value ofits non-inverting input), plus any input offsets, by driving through thenon-inverting input of amplifier 1852. Amplifier 1852 brings itsnon-inverting input to the same potential as VREFL (the value of itsinverting input) plus any input offsets by driving through the invertinginput of amplifier 1851. The output of amplifier 1851 will be at VREFLminus the offset of amplifier 1852. The output of amplifier 1852 will beat VREFL minus the offset of amplifier 1851. Therefore capacitor 1860will have the difference of the offsets of the two amplifiers impressedacross it, effectively zeroing the offsets of the circuit. This is thecondition to which the circuit will return at the conclusion of eachconversion. Note that this configuration is the same for both a four andfive connection arrangement (i.e., FIG. 18A and FIG. 18B).

Referring to FIG. 51G, a conversion to read the position of contact withthe Y coordinate sheet (18.82 in a four-contact implementation) connectsVREFH to TOP (18.83) of the Y coordinate sheet and VREFL to BOT (18.84)of the Y coordinate sheet. The SEL signal is LOW. At the point ofcontact with the X coordinate sheet a voltage is developed which isproportional to the Y coordinate of the point of contact. This voltageis applied to the non-inverting input of amplifier 1851. A high inputimpedance to amplifier 1851 is desirable to provide good accuracy indetermining the contact point. Amplifier 1851 is constructed from fieldeffect transistors which require very little bias current so theresistance of the X coordinate sheet 1881 will not contribute anappreciable error. Amplifier 1851 drives its inverting input to thepotential of its non-inverting input through capacitor 1860. This causesits output initially to equal the voltage present at its non-invertinginput. The level of signal DOUT at the output of amplifier 1852 goes toa high state because its non-inverting input, driven by amplifier 1851,will be above its inverting input which is connected to VREFL. SignalDOUT remains high until the conclusion of the PCA period (FIG. 16). Thevoltage present at the non-inverting input of amplifier 1851 will appearacross resistor 1857 until the PMA period completes. This causes acurrent proportional to position of the contact with the Y coordinatesheet 18.82 to flow into capacitor 1860. The output of amplifier 1851ramps positively to maintain its inverting input at the same potentialas its non-inverting input. At the conclusion of the PMA time, capacitor1860 has a charge proportional to the position of contact with the Ycoordinate sheet 18.82 and the length of time of the PMA phase.

When the active ZERO signal is received into digital controller block1802, QSADC module 1211 a switches into the conversion cycle (PCA) forthe voltage at Port X. At the start of the conversion cycle, up/downcounter 1803 switches to count increment mode (note that the countervalue is already zero). The number of count increments required for theDOUT pin (i.e., output terminal of analog comparator 1852) to return toa low voltage is directly proportional to the ratio of the voltagemeasured at the Port Y to the reference voltage VREFH.

Conversion is completed by setting switch positions as depicted in FIG.51H. The non-inverting input of amplifier 1851 is connected to VREFLthrough switch 1853 e. Amplifier 1851 will drive its inverting input toVREFL through capacitor 1860. This causes VREFL to be applied to oneside of resistor 1857 with VREFH applied to the other through switch1853 k. This causes a current proportional to VREFL minus VREFH acrossresistor 1857 to flow out of capacitor 1860. It should be noted that thecurrent flowing during the PCA phase is the same magnitude (but reversedin sign) as the current flowing during the PMA phase if the point ofcontact is at the end of the sheet connected to VREFH. To the extentthat the point of contact is closer to the end connected to VREFL thePMA current will be proportionately less. The output of amplifier 1851will ramp negatively to maintain its inverting input at the samepotential as its non-inverting input. The PCA phase concludes when theoutput of amplifier 1851 crosses below VREFL, the potential present atthe inverting input of amplifier 1852, at which time DOUT will fall to alow state. The transition of DOUT marks the termination of the PCA. Thelength of time in the PCA phase divided by the length of time in theprevious state will be proportional to the point of contact. When thetimes are equal (i.e., the ratio is one), the point of contact was atthe VREFH end of the sheet. If the ratio is one half, the point ofcontact is half way between the VREFH and VREFL ends of the sheet. Ifthe ratio is zero (time in PCA phase is zero), the point of contact isat the VREFL end of the sheet. After the end of PCB phase has beenrecognized, digital controller block 1802 transfers the count in up/downcounter 1803, which represents a digital value of the voltage at Port X,to register 1805 (REGY), and the circuit again enters the MeasurementInitial State to prepare for the next measurement.

Similarly, referring to FIG. 51I, a conversion to read the position ofcontact with the X coordinate sheet (18.81 in a four-contactimplementation) connects VREFH to RHT (18.86) of the X coordinate sheetand VREFL to LFT (18.85) of the X coordinate sheet. The SEL signal isLOW. At the point of contact with the Y coordinate sheet, a voltage isdeveloped which is proportional to the X coordinate of the point ofcontact. This voltage is applied to the non-inverting input of amplifier1851. The operation of amplifiers 1851 and 1852 is identical to thatdescribed previously for the PMA phase for the Y direction. At theconclusion of the PMB time, capacitor 1860 has a charge proportional tothe position of contact with the X coordinate sheet and the length oftime of the PMB phase.

Similarly, conversion is completed by setting switch positions asdepicted in FIG. 51J. The non-inverting input of amplifier 1851 isconnected to VREFL through switch 1853 e. The operation of amplifiers1851 and 1852 is identical to that described previously for the PCAphase in the Y direction. After the end of PCB phase has beenrecognized, digital controller block 1802 transfers the count in up/downcounter 1803, which represents a digital value of the voltage at Port Y,to register 1804 (REGX), and the circuit again enters the MeasurementInitial State to prepare for the next measurement.

Similarly, referring to FIG. 51K, a conversion to read the verticalposition of contact with the X-Y coordinate sheet (in a five-contactimplementation) connects VREFH to TOP (18.97) of the X-Y coordinatesheet 18.93 and VREFL to BOT (18.98) of the X-Y coordinate sheet(18.93). Note the SEL signal is HIGH. At the point of contact with thecontact sheet 18.92 a voltage is developed which is proportional to theY coordinate of the point of contact. This voltage is applied to thenon-inverting input of amplifier 1851 through switches SEL 1853 p andMEAS 1853 n wherein the contact MID has been connected to connection18.96 of contact sheet 18.92. The operation of amplifiers 1851 and 1852is identical to that described previously for the PMA phase for the Ydirection. At the conclusion of the PMA time, capacitor 1860 has acharge proportional to the position of contact with the Y coordinatesheet 18.93 and the length of time of the PMB phase.

Similarly, conversion is completed by setting switch positions asdepicted in FIG. 51L. Note that SEL is now LOW. The non-inverting inputof amplifier 1851 is connected to VREFL through switch 1853 e. Theoperation of amplifiers 1851 and 1852 is identical to that describedpreviously for the PCA phase in the Y direction. After the end of PCAphase has been recognized, digital controller block 1802 transfers thecount in up/down counter 1803, which represents a digital value of thevoltage at the MID port, to register 1805 (REGY), and the circuit againenters the Measurement Initial State to prepare for the nextmeasurement.

Referring to FIG. 51M, a conversion to read the horizontal position ofcontact with the X-Y coordinate sheet (in a five-contact implementation)connects VREFH to LHT (18.94) of the X-Y coordinate sheet and VREFL toRHT (18.95) of the X-Y coordinate sheet. Note the SEL signal is HIGH. Atthe point of contact with the contact sheet 18.92 a voltage is developedwhich is proportional to the X coordinate of the point of contact. Thisvoltage is applied to the non-inverting input of amplifier 1851 throughswitches SEL 1853 p and MEAS 1853 n wherein the contact MID has beenconnected to connection 18.96 of contact sheet 18.92. The operation ofamplifiers 1851 and 1852 is identical to that described previously forthe PMA phase for the Y direction. At the conclusion of the PMA time,capacitor 1860 has a charge proportional to the position of contact withthe X coordinate sheet and the length of time of the PMB phase

Similarly, conversion is completed by setting switch positions asdepicted in FIG. 51N. Signal SEL is now LOW. The non-inverting input ofamplifier 1851 is connected to VREFL through switch 1853 e. Theoperation of amplifiers 1851 and 1852 is identical to that describedpreviously for the PCA phase in the Y direction. After the end of PCBphase has been recognized, digital controller block 1802 transfers thecount in up/down counter 1803, which represents a digital value of thevoltage at the MID port, to register 1804 (REGX), and the circuit againenters the Measurement Initial State to prepare for the nextmeasurement.

FIG. 51 is a block diagram showing interface signals of QSADC module1211 b in another implementation. The implementation of FIG. 51 issimilar to the implementation in FIG. 17 and therefore the discussionfor similar features will not be repeated. Table 1 provides a briefdescriptive summary of the signals shown in FIG. 51.

The implementation of FIG. 51 includes a digital interface 1902.4 thatis different than digital interface 1702 (FIG. 17). Specifically,digital interface 1902.4 comprises one 8-bit output bus 1902 a.4(labeled DOUT[7:0]) and a Select signal 1902 b.4 (labeled SEL), ratherthan output buses 1702 a and 1702 b as described in reference to FIG.17. Output bus 1902 a.4 provides the results of the analog-to-digitalconversion (ADC) process of the voltage at the port A or the port B withrespect to the voltage reference VREF.

Select signal 1902 b.4 is the output select signal that selects whetherto present the results from the port A or the port B after the DONEsignal is asserted. For example, if SEL 1902 b.4 is a logical LOW level,then the ADC result of the voltage at the port A is presented on outputbus 1902 a.4. Otherwise, if select signal 1902 b.4 is a logical highlevel, then the ADC result of the voltage at the port B is presented onoutput bus 1902 a.4.

TABLE 1 Signal/Pin Description Pin Name Type Description AMAXBidirectional Maximum Analog Measurement Pin for the A Port AMINBidirectional Minimum Analog Measurement Pin for the A Port BMAXBidirectional Maximum Analog Measurement Pin for the B Port BMINBidirectional Minimum Analog Measurement Pin for the B Port START InputStart Conversion Pin Active HIGH CLK Input Clock RSTN Input Reset ActiveLOW PDN Input Power Down Control Active LOW DOUT[7:0] Output DigitizedBus for Measured Value DONE Output Done Pin Active HIGH CONT OutputContinuity Detection Pin Active HIGH VREF Input Analog Voltage ReferenceAVD Power Analog Power AVS Power Analog Ground VDD Power Digital PowerVSS Power Digital Ground

FIG. 51A and FIG. 51B show another implementation of QSADC module 1211b, in accordance with the present invention. FIG. 51A and FIG. 51B aresimilar to FIG. 18 and FIG. 18A and operate in a similar fashion and,therefore, only basic operational differences between the embodimentswill be specifically noted.

FIG. 51A is a top-level block diagram of QSADC module 1211 b, includingan analog block 2001.4, a digital controller block 2002.4, apre-settable up/down counter block 2003.4, registers 2004.4 and 2005.4,a NAND gate 2006.4, and a multiplexer 2007.4. FIG. 51A differs from FIG.18 due mainly to the addition of select signal 1902 b.4 (labeled SEL) inFIG. 51A. Select signal 1902 b.4 employs multiplexer 2007.4 to selectthe data stored in register 2004.4 or register 2005.4 to be provided onoutput bus 1902 a.4 (DOUT[7:0]). For example, the digital conversionresults of the voltages at the port A and the port B are stored inregister 2004.4 and register 2005.4, respectively. If SEL 1902 b.4 is ata logical low level, the conversion results of the port A (stored inregister 2004.4) are provided on output bus 1902 a.4 via multiplexer2007.4. If SEL 1902 b.4 is at a logical high level, the conversionresults of the port B (stored in register 2005.4) are provided on outputbus 1902 a.4 (DOUT[7:0]) via multiplexer 2007.4.

FIG. 51B shows an implementation of analog block 2001.4 of FIG. 51A. Theoperation of the implementation shown in FIG. 51B is similar to theoperation of the implementation shown in FIG. 18A and, therefore, thediscussion will not be repeated except for noting general differences.

As shown in FIG. 51B, analog block 2001.4 includes a digital controlblock (DCNTL) 2020.4 and a level shift circuit 2022.4. Initially, QSADCmodule 1211 b is in sleep mode, but checking for electrical continuitybetween the ports A (AMAX and AMIN) and B (BMAX and BMIN), with a sleepcontrol signal (SLP) at a logic high, a power down signal (PDON) at alogic low, and operational amplifier 1851 and analog comparator 1852powered down to conserve power. In this mode of operation, switches 1853c, 1853 b, 1853 h, 1853 i, 18531, 1853 n, and 1853 f (i.e., signals A0,A2, B1, B2, MEAS, and SLP) are closed, while all other switches areopen.

When there is no electrical continuity between the ports Y and X, acontinuity out (CONT) signal is at a logical low level. When electricalcontinuity is established, the input to level shift circuit 2002.4 ispulled to a high voltage (i.e., to a voltage level of the voltagereference) and the continuity out signal transitions to a logical highlevel. The logical high on the continuity out signal is detected (andde-bounced) by a flip flop (not shown) in digital controller block2002.4, resulting in digital controller block 2002.4 asserting a logicalhigh signal on the CONT terminal. QSADC module 1211 b then exits thesleep mode and proceeds to power up operational amplifier 1851 andanalog comparator 1852 to begin the primary mode of operation (asdiscussed above). During the primary mode, the circuitry associated withthe continuity checking in digital controller block 2002.4 may bedisabled.

FIG. 51C illustrates a block diagram showing exemplary interface signalsfor another implementation. The block diagram illustrates a digitallogic wrapper mixed-signal macro (also referred to as a touch screeninterface or TSI) that encapsulates QSADC 1211 a. The wrapper providesan interface to the 8051 micro-controller in central processing module(SYS) 1205 for reading the touch screen coordinate data from QSADC 1211a.

The wrapper contains data, control, and status registers, as describedin further detail below, that allow the software driver to work moreefficiently by presenting the necessary information in specificlocations. An internal state machine in the wrapper will manage theoperation of QSADC 1211 a and its interaction with touch screeninterface 1211.

Touch screen interface 1211 uses the three interfaces AIO 1701, powersupply interface (PSI) 1703, and DI 1702. AIO 1701 and power supplyinterface 1703 are as described above, while DI 1702 also includessignals required by central processing module (SYS) 1205 to control theoperation of QSADC 1211 a and access the sampled data. These signalsinclude a SYS_CLK signal 5002.4, a TS_SFR_REG_EN signal 5010.4, aSFR_ADDR[7:0] signal 5012.4, a SFR_DATA_OUT[7:0] signal 5014.4, aSFR_WR_N signal 5016.4, a SFR_RD_N signal 5018.4, a TS_SFR_DATA_IN[7:0]signal 5042.4, a TS_MEM_REG_EN signal 5020.4, a MEM_ADDR[2:0] signal5022.4, a MEM_DATA_OUT[7:0] signal 5024.4, a MEM_WR_N signal 5026.4, aMEM_RD_N signal 5028.4, a TS_MEM_DATA_IN[7:0] signal 5040.4, a TS_INTsignal 5044.4, and a QSADC_CLK signal 5004.4.

RSTN signal 5000.4 is the chip-level reset used to initialize theinternal logic of the TSI. SYS_CLK signal 5002.4 is the clock used tosynchronize the TSI's internal logic to central processing module (SYS)1205. QSADC_CLK signal 5004.4 is the clock used to control touch screeninterface 1211 state machine and QSADC 1211 a. TS_SFR_REG_EN signal5010.4 indicates that touch screen interface 1211 register on the SFRbus, a component of central processing module (SYS) 1205, is about to beaccessed. The SFR bus enables access to non-memory mapped registers, asdescribed elsewhere herein.

SFR_ADDR[7:0] signal 5012.4 is the address received from the 8051 usedto access one of the SFR registers. SFR_DATA_OUT[7:0] signal 5014.4contains the data (i.e. control word) to be written into the registers.SFR_WR_N signal 5016.4 is used in conjunction with TS_SFR_REG_EN signal5010.4 to write TS_SFR_DATA_IN[7:0] signal 5042.4 to one of theregisters addressed by SFR_ADDR[7:0] signal 5012.4. SFR_RD_N signal5018.4 is used in conjunction with TS_SFR_REG_EN signal 5010.4 to readSFR_DATA_OUT[7:0] signal 5014.4 from one of the registers addressed bySFR_ADDR[7:0] signal 5012.4. TS_SFR_DATA_IN[7:0] signal 5042.4 containsthe data (i.e. data/status words) to be read from the registers.

TS_MEM_REG_EN signal 5020.4 indicates that a TSI register on the MemoryMapped I/O (MMIO) bus, a component of central processing module (SYS)1205 is about to be accessed. MEM_ADDR[2:0] signal 5022.4 is the addresscoming from the 8051 used to access one of the MMIO registers. Thisaddress bus is a partial decode of the 16-bit MEM_ADDR[2:0] signal5022.4 on the 8051 (specifically bits [2:0]). All MMIO registertransactions are qualified with the TS_MEM_REG_EN signal 5020.4. Using apartial decode reduces routing congestion and eliminates the need for afull decode.

MEM_DATA_OUT[7:0] signal 5024.4 contains the data (i.e. control word) tobe written into the registers. MEM_WR_N signal 5026.4 is used inconjunction with TS_MEM_REG_EN signal 5020.4 to writeTS_MEM_DATA_IN[7:0] signal 5040.4 data to one of the registers addressedby MEM_ADDR[2:0] signal 5022.4. MEM_RD_N signal 5028.4 is used inconjunction with TS_MEM_REG_EN signal 5020.4 to read MEM_DATA_OUT[7:0]signal 5024.4 from one of the registers addressed by MEM_ADDR[2:0]signal 5022.4. TS_MEM_DATA_IN[7:0] signal 5040.4 contains the data (i.e.data/status words) to be read from the registers.

TS_INT signal 5044.4 is an interrupt line sent to the 8051 containedwithin central processing module (SYS) 1205 to indicate that an (X, Y)coordinate pair has been converted to a digital value by QSADC 1211 aand is ready for use. The 8051 treats this line as an edge-sensitiveinterrupt, with the line remaining high until the 8051 clears it.

TABLE 1.6.6a Signal Type Description Source Destination RSTN InputGlobal chip reset QSADC_CLK Input QSADC 1211a macro clock CLKGEN SYS_CLKInput SYS 1205 module clock CLKGEN AIO Interface AIO PSI Interface AIOTS_SFR_REG_EN Input Enables touch screen SFR SYS 1205 registerSFR_ADDR[7:0] Input SFR address from 8051 SYS 1205 SFR_DATA_OUT[7:0]Input SFR data from 8051 SYS 1205 SER_WR_N Input SFR data write enableSYS 1205 SFR_RD_N Input SFR data read enable SYS 1205TS_SFR_DATA_IN[7:0] Output SFR data to 8051 SYS 1205 TS_MEM_REG_EN InputEnable touch screen MEM SYS 1205 register MEM_ADDR[2:0] Input MMIOaddress from 8051 SYS 1205 MEM_DATA_OUT[7:0] Input MMIO data from 8051SYS 1205 MEM_WR_N Input MMIO data write enable SYS 1205 MEM_RD_N InputMMIO data read enable SYS 1205 TS_MEM_DATA_IN[7:0] Output MMIO data to8051 SYS 1205 TS_INT Output Interrupt to 8051 SYS 1205 indicating an (X,Y) coordinate pair is ready for reading.

FIG. 51D shows a flowchart for performing diagnostics for the TSI andFIG. 51E shows a functional block diagram corresponding to the TSI. InFIG. 51E, besides QSADC 1211 a, various registers are shown, including aTSI_CTRL[7:0] register 5062.4, an SMP_DELAY[7:0] register 5060.4, anINT_STAT[7:0] register 5064.4, an X_DATA[7:0] register 5066.4, and aY_DATA[7:0] register 5068.4.

Specifically, TST_CTRL[7:0] 5062.4 controls the operation of touchscreen interface 1211 and QSADC 1211 a. Bit 0 of TSI_CTRL[7:0] 5062.4enables or disables the TSI. If this register bit is ‘0’, an internalstate machine (FSM) 5070.4 stays in IDLE and QSADC 1211 a is inlow-power mode. If this register bit is ‘1’, the state machine startscollecting samples as soon as it detects activity on the touch screen.Bit 1 of TSI_CTRL[7:0] 5062.4 indicates the type of touch screen that isattached to the Power Meister (E.G., 4-pin or 5-pin).

SMP_DELAY[7:0] register 5060.4 controls the time between samples. Forexample, QSADC 1211 a is capable of collecting a pair of (X, Y)coordinate samples every 3.9 ms given a 262.144 kHz clock. However, thisrate may be too fast for certain operating systems to handle, soSMP_DELAY[7:0] register 5060.4 allows the TSI to vary the waiting timebetween successive samples from 0 ms to 6.4 ms. The maximum samplecollection rate would then be 3.9 ms to 10.3 ms respectively.

INT_STAT[7:0] register 5064.4 is the interrupt status register. When aninterrupt is detected, TSI_INT signal 5044.4 is held high until the 8051clears the particular interrupt.

A DIAG_CTRL[7:0] register (not shown) controls the diagnostic logic andis routed directly to the QSADC 1211 a block via the MMIO bus. In thismode, internal state machine 5070.4 is bypassed and QSADC 1211 a isdirectly under software control. The 8051 will set the DIAG_CTRL[1] bitto ‘1’ to reset the logic and the TSI will then clear this bit. When the8051 sets the DIAG_CTRL[2] bit to ‘1’, QSADC 1211 a retrieves 1 samplepair. The 8051 polls the DIAG_CTRL[5] bit to determine when a sample isavailable for reading.

X_DATA[7:0] register 5066.4 contains X coordinate data from the touchscreen. QSADC 1211 a latches X coordinate data into this register andasserts DONE signal 1702 c, which is then sent as an interrupt line tocentral processing module (SYS) 1205 to allow the software driver toread the data.

Y_DATA[7:0] register 5068.4 contains Y coordinate data from the touchscreen. QSADC 1211 a latches Y coordinate data into this register andasserts DONE signal 1702 c, which is then sent as an interrupt line tocentral processing module (SYS) 1205 to allow the software driver toread the data. Table 1.6.6b summarizes information for the variousregisters.

TABLE 1.6.6b Register Name Address Type Addressing DescriptionTSI_CTRL[7:0] R/W MMIO Controls the operation of the TSI [7:6] -undefined [1] - touch screen mode select 0 = 4-pin 1 = 5-pin [0] - TSIenable (0 - disable, 1 - enable) SMP_DELAY[7:0] R/W MMIO Sample delay[7:6] - undefined [5:0] - sample delay in increments of 100 us (0x00 =no delay, 0x3F = 6.4 ms) INT_STAT[7:0] R/W SFR Interrupt Status Register[7:6] - undefined [5] - conversion underrun error [4] - conversionoverrun error [3] - undefined [2] - data ready [1] - pen down [0] - penup DIAG_CTRL[7:0] R/W MMIO Diagnostic Control and Status [7:6] -undefined [5] - data ready [4] - QSADC ready [3] - undefined [2] - startsingle conversion [1] - reset TSI (set by 8051, cleared by TSI) [0] -diagnostic mode enable X_DATA[7:0] R SFR 8-bit X coordinate data fromthe touch screen Y_DATA[7:0] R SFR 8-bit Y coordinate data from thetouch screen

In terms of general operation, the 8051 will set bit 0 (i.e.,TSI_CTRL[0]) of TSI_CTRL[7:0] register 5062.4 to ‘1’ to enable the TSIafter the chip (i.e., switching power supply controller 1200) has beenpowered up and the global chip reset has been asserted. When touchscreen interface 1211 is enabled, internal state machine 5070.4 willretrieve (X, Y) sample pairs as long as QSADC 1211 a has detectedcontinuity (i.e. a pen-down condition exists). After QSADC 1211 a hascollected a sample pair, it will wait for a specified amount of time, asindicated by SMP_DELAY[7:0] register 5060.4, before starting over andcollecting the next sample pair. This cycle continues as long ascontinuity is detected by QSADC 1211 a.

Internal state machine (FSM) 5070.4 schedules the conversion of touchscreen data by toggling START/RSTN signal 1702 e of QSADC 1211 a andthen sampling DONE signal 1702 c to load X/Y data registers 5066.4 and5068.4. The sampling will occur periodically as long as CONT signal 1702d of QSADC 1211 a is asserted.

Every time a sample is collected or a pen-down/pen-up condition isdetected (as seen by a toggling of the CONT 1702 d signal), an interruptis sent to the 8051. This interrupt line is held high until the 8051clears the corresponding bit in INT_STAT[7:0] register 5064.4. Aninterrupt is also sent if a buffer overrun/underrun condition occurs.The TSI doublebuffers the X/Y coordinate samples and the activity onthese buffers is tracked to generate the appropriate interrupt.

Referring to FIG. 51D and FIG. 51E, when the TSI is placed in diagnosticmode by setting DIAG_CTRL[4] of DIAG_CTRL[7:0] register 5062.4 to ‘1’,FSM 5070.4 is bypassed and START/RSTN signal 1702 e of QSADC 1211 a isdirectly controlled by setting bit 5 of DIAG_CTRL[7:0] register 5062.4.This mode of operation is intended for use as a debug mechanism toverify the operation of the TSI and QSADC 1211 a. Interrupts to the 8051are disabled and a polling mechanism is used where bit 1 (i.e.,INT_STAT[1]) of INT_STAT[7:0] register 5064.4 acts as DONE signal 1702c.

The test routine shown in FIG. 51D is as follows. The 8051 (step 5050.4)reads DIAG_CTRL[4] to ensure QSADC 1211 a is ready. If QSADC 1211 a isnot ready, the 8051 must first assert DIAG_CTRL[1] to reset touch screeninterface 1211 and QSADC 1211 a. The 8051 sets DIAG_CTRL[2] to ‘1’, thenthe 8051 polls DIAG_CTRL[5] until TRUE. When TRUE, 8051 reads the values(5054.4) in X_DATA[7:0] register 5066.4 and Y_DATA[7:0] register 5068.4.Following a WAIT period (5056.4), the flow returns to IDLE (step 5058.4)and repeats as necessary.,

Several advantages compared to the prior art may be seen. The inventioneliminates sensitivity to noise. The most difficult noise source in aPDA application is the high voltage, high frequency a.c. signal thatdrives the CCFL or other type of backlight. By arranging clockfrequencies such that there is always an even number of backlight pulsesduring a conversion period (e.g., 1601+1602 of FIG. 16) any noise thatis picked up from one backlight half cycle is subtracted back out by thenext half cycle. Since all clocks are controlled by switching powersupply controller 1200, this may be arranged. In addition, the systemdoes not drift due to component aging or thermal effects nor does itrequire calibration, either at the time of manufacture or later. Also,the invention does not require precision components (typically resistorsand capacitors) in that the same resistor and capacitor are used duringthe measurement and the conversion phases. Thus any “errors” incomponent value during 1601 or 1602 times (FIG. 16) are reversed during1605 and 1606 times.

Quad-Slope Analog to Digital Converter (QSADC) module 1211 includes acustom low-power mixed-signal circuit that has both analog and digitalinput and output signals, including analog and digital power and groundreference voltages. QSADC module 1211 measures and digitizes, to 8 bitsof resolution, the voltages at two separate external ports relative to avoltage reference VREF. The maximum conversion rate at QSADC module 1211is approximately 300 sps (samples per second). FIG. 16 is a timingdiagram illustrating the quad-slope (i.e., dual conversion) analog todigital conversion (ADC) operations carried out in the QSADC module1211.

As shown in FIG. 16, time intervals 1601 and 1602 (i.e., “measurementperiods” PMA and PMB) are equivalent fixed time durations correspondingto 256 counts of a reference timer or counter. During time periods 1601and 1602, the positive slopes if signal trace segments 1603 and 1604reaches voltages V_(A) and V_(B), represent signals being measured atthe analog A and B ports of the QSADC module 1211 over time periods 1601and 1602. During time intervals 1605 and 1606 (“conversion periods” PCAand PCB), signal trace segment 1607 and 1608 have identical negativeslopes, tracing in voltage declines from voltages V_(A) and V_(B) tozero. During these conversion periods, the numerical counts representdirect analog to digital conversions of the measured voltages V_(A) andV_(B). At the completions of the conversion periods, end-of-conversion(EOC) pulses are provided (e.g., at times t₁ and t₂) to signal the endof each analog to digital conversion and to reset and calibrate theanalog circuitry in QSADC module 1211 for subsequent conversions.

FIG. 17 is a block diagram showing the interface signals of QSADC module1211 in one implementation. As shown in FIG. 17, QSADC module 1211includes: (a) analog measurement interface (AMI) 1701, digital interface(DI) 1702, and the power supply interface 1703. Analog measurementinterface (AMI) comprises analog bi-directional measurement terminals1701 a and 1701 b of port A (i.e., terminals AMAX and AMIN), analogbi-directional measurement terminals 1701 c and 1701 d of port B (i.e.,terminals BMAX and BMIN) and reference voltage signal 1701 e (i.e.,reference voltage VREF).

Analog measurement interface 1701 operates in two modes in each terminalof each port. In a first mode (“primary mode”), terminal 1701 a (AMAX)measures two independent external voltages relative to voltage referenceVREF at terminal 1701 e. The primary mode includes two phases. Duringthe first phase, terminals 1701 a and 1701 b (i.e., AMAX and AMINterminals) are shorted to allow port A to float and to complete thefirst digital voltage conversion (at the port A). During the secondphase, terminal 1701 a (i.e., terminal AMAX) is connected to terminal1701 e, which carries reference voltage VREF, to facilitate the seconddigital voltage conversion (at port B). In a second mode (“low power”)terminals 1701 a and 1701 b (i.e., AMAX and AMIN terminals) are againshorted to allow port A to float. During low power, port A and port Boperate to check for electrical continuity between these ports.

In the primary mode, terminal 1701 b measures two independent externalvoltages with respect to a voltage reference, VREF. In the primary mode,the first phase for terminal 1701 b is shared with the correspondingfirst phase for terminal 1701 a, as described above. In the second phaseof the primary mode, terminal 1701 b connects to analog ground reference(AVS) at terminal 1703 d to facilitate the second digital voltageconversion (at port B). The operations of terminal 1701 b's low powerare the same as those of terminal 1701 a under low power.

Similarly, in the primary mode, terminal 1701 c (BMAX) measures twoindependent external voltages relative to voltage reference VREF atterminal 1701 e. During a first phase, terminals 1701 c and 1701 d(i.e., BMAX and BMIN terminals) are shorted to allow port B to float andto complete the second digital voltage conversion (at the port B).During the second phase, terminal 1701 c (i.e., terminal BMAX) isconnected to terminal 1701 e, which carries reference voltage VREF, tofacilitate the second digital voltage conversion (at port A). In asecond mode (“low power”) terminals 1701 c and 1701 d (i.e., BMAX andBMIN terminals) are again shorted to allow port B to float. During lowpower, port A and port B operate to check for electrical continuitybetween these ports.

In the primary mode, terminal 1701 d measures two independent externalvoltages with respect to a voltage reference, VREF. In the primary mode,the first phase for terminal 1701 d is shared with the correspondingfirst phase for terminal 1701 c, as described above. In the second phaseof the primary mode, terminal 1701 d connects to analog ground reference(AVS) at terminal 1703 d to facilitate the first digital voltageconversion (at port A). The operations of terminal 1701 d's low powerare the same as those of terminal 1701 c under low power.

Digital interface 1702 comprises 8-bit output buses 1702 a (AOUT[7:0])and 1702 b (BOUT[7:0]) for displaying digital results of the conversionsat ports A and B, respectively, completion or “DONE” signal 1702 c,continuation detection or “CONT” signal 1702 d, start conversion or“START” signal 1702 e, 128 KHz 50% duty cycle reference clock signal(“CLK”) 1702 f, asynchronous reset signal (“RSTN”) 1702 g, andpower-down signal (“PDN”) 1702 h. START is the start control pin used toinitiate the analog to digital conversion processes. When the START pingoes HIGH the processing starts and subsequent conversions will not beinitiated until the START pin toggles LOW then HIGH again.

The DONE signal at terminal 1702 c is asserted when the digitalconversions at both ports A and B are completed, to signal that resultscan be read from output buses 1702 a and 1702 b (i.e., AOUT and BOUTbuses). The CONT signal indicates detection of continuity at ports A andB.

Power supply interface 1703 comprises analog power and ground referencesignals 1703 a and 1703 b (i.e., analog power and ground referencesignals AVD and AVS), and digital power and ground reference signals1703 c and 1703 d (i.e., digital power and ground reference signals VDDand VSS).

FIG. 18 and FIG. 18A show one implementation of QSADC module 1211, inaccordance with the present invention. FIG. 18 is a top-level blockdiagram of QSADC module 1211, including analog block 1801, digitalcontroller block 1802, presettable up/down counter block 1803, 8-inputNAND gate 1806, and 8-bit Registers 1804 and 1805 (i.e., registers REGAand REGB). FIG. 18A shows one implementation of analog block 1801 ofFIG. 18. As shown in FIG. 18A, analog block 1801 includes operationalamplifier 1851 and analog comparator 1852, in addition to MOStransmission gates 1853 a to 1853 m, inverter 1854 and digital controlblock (DCNTL) 1855. DCNTL block 1855 generates the control signalsnecessary to control the MOS transmission gates 1853 a to 1853 m. Thecommon mode ranges of the operational amplifier 1851 and analogcomparator 1852 are rail to rail (i.e., 0 to 3.3V). For convenience theMOS transmission gates 1853 a to 1853 m are referred to as “switches”and each switch is designated by its control input signal (i.e., A0-A3,B0-B3, MEAS, SLP, EOC, CONV). Initially, QSADC module 1211 is in lowpower, where electrical continuity between port A (i.e., terminals 1701a (AMAX) and 1701 b (AMIN)) and port B (i.e., terminals 1701 c (BMAX)and 1701 d (BMIN)) is checked. Under low power, switches 1853 b, 1853 c,1853 h and 1853 i, 1853 d, 18531, and 1853 f (i.e., signals A0, A2, B1,B2, MEAS and SLP) are closed, while all other switches are open. Whenthere is no electrical continuity between ports A and B, the outputvoltage of operational amplifier 1851 is a low voltage, and the outputvoltage of analog comparator 1852 is also a low voltage. When electricalcontinuity is established between ports A and B, the output terminal ofswitch 1853 d (i.e., the non-inverting input terminal of operationalamplifier 1851) is pulled to a high voltage, so that the output terminalof analog comparator 1852 is at a high voltage. The output terminal ofanalog comparator 1852 is the “DOUT” terminal of analog block 1801 (FIG.18), which is connected to the “DIN” terminal of digital control block1802. In response to the high voltage at the output terminal of analogblock 1801, the CONT terminal (i.e., terminal 1702 d at FIG. 17) isdriven to a high voltage, resulting in QSADC module 1211 exiting the lowpower, and entering the primary mode.

In primary mode, entered when the CONT signal is at a high voltage, whenan active START signal at terminal 1702 e is received, digital controlblock 1802 begins a measurement cycle of the voltage at port A bypresetting up/down counter 1803 to hexadecimal FF (i.e., FFh), therebyresetting the CONT signal, and beginning the measurement and conversionprocess. Thereafter, each rising edge of the CLK signal at terminal 1702f decrements up/down counter 1803 until the count reaches zero (for atotal of 256 counts), which is decoded by NAND gate 1806 to activate theZERO signal received into digital control block 1802.

During the measurement cycle of port A, switches 1853 b, 1853 c, 1853 hand 1853 i, 1853 d, 18531, and 1853 f (i.e., signals A0, A2, B1, B2,MEAS) are closed, and all other switches are open. As operationalamplifier 1851 is configured as an integrator with capacitor 1856, aspecific charge approximately proportional to the voltage at port A isaccumulated across capacitor 1856 over the time interval correspondingto 256 count decrements of up/down counter 1803.

When the active ZERO signal is received into digital controller block1802, QSADC module 1211 switches into the conversion cycle for thevoltage at port A. At the start of the conversion cycle, up/down counter1803 switches to count increment mode. During the conversion cycle, theswitches 1853 e and 1853 k controlled by the CONV signal are closed,while all other switches are open, so that the charge on capacitor 1856discharges reference resistor 1857 (R0). Therefore, the number of countdecrements required for the DOUT pin (i.e., output terminal of analogcomparator 1852) to return to a low voltage is directly proportional tothe ratio of the voltage measured at the port A to the reference voltageVREF. The conversion cycle terminates when the high to low voltagetransition occurs at analog comparator 1852, and detected by digitalcontroller block 1802 at the terminal carrying signal DIN. At this time,digital controller block 1852 transfers the count in up/down counter1803, which represents a digital value of the voltage at port A, toregister 1804. At this time, digital controller 1802 closes and opensswitch 1853 n via control signal EOC, to compensate any voltage offsetacross capacitor 1856, so as to ready QSADC module 1211 for a subsequentport B measurement and conversion cycle. The measurement and conversioncycles for port B are substantially the same as that described above forthe port A. The result of a port B conversion cycle is stored inregister 1805 (REGB). The DONE signal (terminal 1702 c) is then assertedand QSADC module 1211 returns to sleep mode, until the next assertedSTART signal (terminal 1702 e) is received, or when electricalcontinuity is detected across ports A and B.

While several embodiments of this invention have been shown, otherembodiments of this invention will be obvious to those skilled in theswitching power supply design arts.

1. A triggering unit for triggering state changes in pulse widthmodulated output signals of a plurality of pulse width modulatingchannels, the triggering unit comprising: reprogrammable memory forstoring first data words which can be used to determine, on a perchannel basis, the timings of one or both of a beginning and an end of arespective pulse in a respective one of said pulse width modulatedoutput signals; and a comparator, operatively coupled to the memory,adapted to compare a supplied, timing reference signal to second datathat is derived from one or more of said first data words provided bythe memory and providing a change-triggering output signal fortriggering a change of state in a corresponding pulse width modulatedoutput signal in response to a predetermined comparison conditionbetween the timing reference signal and the second data of thecorresponding channel.
 2. The triggering unit of claim 1, wherein thememory includes one or more, multi-ported random access memory unitseach having independently addressable, data input and data output ports.3. The triggering unit of claim 1, further comprising data modifyingmeans, coupled to the memory and to the comparator, for modifying thefirst data words obtained from the memory to thereby generate the seconddata prior to receipt by the comparator of the second data.
 4. Thetriggering unit of claim 3 wherein the data modifying means includes anadder and an accumulator, the accumulator being operatively coupled toreceive summation result signals produced by the adder and to supply thesummation result signals as next addends to an addend input port of theadder.
 5. The triggering unit of claim 1 wherein the memory hassufficient storage capacity to store first data words for at least sevenpulse width modulating channels.
 6. The triggering unit of claim 1wherein the supplied, timing reference signal has at least 10 bits oftiming resolution.
 7. A machine-implemented triggering method fortriggering state changes in pulse width modulated output signals of aplurality of activatable pulse width modulating channels, the methodcomprising: (a) generating a reference count signal that is common tosaid plurality of pulse width modulating channels, where the commonreference count signal counts through a sequence of successive timingvalues; (b) for each active channel, generating at least one triggervalue signal that is to be compared against the successive timing valuesof the common reference count signal; (c) for each given, activechannel, comparing the common reference count signal against the atleast one trigger value signal of the given channel, and uponsatisfaction of a predefined comparison condition between comparedvalues of the common reference count signal and the at least one triggervalue signal, outputting a respective comparison output signal for thegiven channel, where the respective comparison output signal is capableof invoking a change in the pulse width modulated output signal of thegiven channel.
 8. The machine-implemented triggering method of claim 7wherein at least two of said plurality of pulse width modulatingchannels are independently activatable such that each can be controlledin response to at least, changed conditions of its corresponding, pulsewidth modulated load.
 9. The machine-implemented triggering method ofclaim 8 wherein the changed condition of the channel's corresponding,pulse width modulated load include at least one of a correspondingchanged voltage and a corresponding changed current draw.
 10. Themachine-implemented triggering method of claim 8 wherein detection of achanged condition of the channel's corresponding, pulse width modulatedload is carried out by an included sampling means whose sampling timesare scheduled relative to said common reference count signal.
 11. Themachine-implemented triggering method of claim 7 wherein said step ofgenerating at least one trigger value signal for each active givenchannel comprises: (b.1) generating at least first and second triggervalue signals for the given channel where the first trigger value signalis usable for determining a turn-ON triggering condition for the pulsewidth modulated output signal of the given channel and the secondtrigger value signal is usable for determining a turn-OFF triggeringcondition for the pulse width modulated output signal of the givenchannel.
 12. The machine-implemented triggering method of claim 7wherein said step of generating at least one trigger value signal foreach active given channel comprises: (b.1) for each active givenchannel, storing trigger-precursor data in a reprogrammable memory; and(b.2) for each active given channel, automatically first computing acorresponding one or more first-computed values for the at least onetrigger value signal of the given channel from the storedtrigger-precursor data.
 13. The machine-implemented triggering method ofclaim 12 wherein said step of generating at least one trigger valuesignal for each active given channel further comprises: (b.3) for eachactive given channel, automatically re-computing a corresponding one ormore of the first-computed values for the at least one trigger valuesignal of the given channel if the first computing produces one or morevalue collisions between trigger values of two or more of said pulsewidth modulating channels, said re-computing constituting a reschedulingof pulse edges so that pulse edges of different channels will not occurat a substantially same time so as to cause cross-effects betweensimultaneously changing channels.
 14. The machine-implemented triggeringmethod of claim 12 and further comprising: (d) for each active givenchannel, and after said first computing of the corresponding one or morefirst-computed values for the at least one trigger value signal of thegiven channel, scheduling one or more measurement samplings to occur attimes displaced along said common reference count signal away from thescheduled state change times for the pulse width modulated output signalof the given channel, said one or more measurement samplings being formeasuring one or more attributes of the pulse width modulated outputsignal of the given channel.
 15. The machine-implemented triggeringmethod of claim 12 wherein said step of first computing includes: (b.2a)determining a difference between a target output voltage to be producedby the pulse width modulated output signal of the given channel and ameasured output voltage produced by the pulse width modulated outputsignal of the given channel; and (b.2b) incrementing or decrementing astored trigger value for the given channel in response to the determineddifference.
 16. The machine-implemented triggering method of claim 7 andfurther comprising: (d) for each given channel, associating a tag signalwith the corresponding at least one trigger value signal, said tagsignal indicating at least which channel, if any, the at least onetrigger value signal corresponds to.
 17. The machine-implementedtriggering method of claim 16 wherein each active channel has at leasttwo tag signals and a corresponding at least two trigger value signalsstored and associated with that active channel.
 18. A sample and controlmethod for use in a pulse-driven power converter, where said powerconverter has a converter output voltage that, when applied to a load,is prone to include an output ripple voltage portion that is a result ofthe pulsed-driving of the power converter, where said power converterincludes an inductor subjected to inductor-driving voltage pulses eachhaving pulse waveform features including a leading edge and a trailingedge, the inductor developing an inductor current in response to saidinductor-driving voltage pulses, where said power converter includes ancapacitor coupled to receive at least part of said inductor current andto convert the received inductor current into said converter outputvoltage; the method comprising: (a) programmably scheduling firsttimings for leading edges of an inductor-driving first voltage pulse tooccur in synchronism with a reference count signal that is common to atleast one other pulse-driven power converter, where the common referencecount signal counts through a sequence of successive timing values; (b)programmably scheduling second timings for obtaining first samples ofthe converter output voltage by sampling the converter output voltage atsecond timing points synchronized to the reference count signal, tothereby substantially exclude from the first samples, effects of theoutput ripple voltage portion.
 19. A triggering circuit for triggeringstate changes in pulse width modulated output signals of a plurality ofactivatable pulse width modulating channels, the circuit comprising: (a)a reference count generator that generates a reference count signalwhich is commonly distributed to said plurality of pulse widthmodulating channels, where the common reference count signal countsthrough a sequence of successive timing values; (b) a trigger valuegenerator that generates at least one respective trigger value signalfor each respective and active channel, where the generated triggervalue signal is to be compared against the successive timing values ofthe common reference count signal; (c) a comparator that compares thecommon reference count signal against the at least one, generated andrespective trigger value signal of each given, active channel, and thatoutputs a respective comparison output signal for the given channel uponsatisfaction of a predefined comparison condition between comparedvalues of the common reference count signal and the at least onerespective trigger value signal, where the respective comparison outputsignal is capable of invoking a change in the pulse width modulatedoutput signal of the given channel.
 20. The triggering circuit of claim19 wherein said trigger value generator includes: (b.1) random accessmemory for storing initial data for defining the at least one respectivetrigger value signal for each respective and active channel; and (b.2)data processing means, operatively coupled to the random access memory,for producing the generated trigger value signals for each respectiveand active channel.
 21. The triggering circuit of claim 19 wherein:(b.2a) said data processing means includes means for incrementing ordecrementing the values of intermediate trigger value signals of given,active channels so as to avoid simultaneous state changes in the pulsewidth modulated output signals of two or more of the active pulse widthmodulating channels.
 22. The triggering circuit of claim 19 wherein:(b.2a) said data processing means includes means for incrementing ordecrementing the values of intermediate trigger value signals of agiven, active channel so as to thereby modify the duty cycle of thepulse width modulated output signal of the given, active channel. 23.The triggering circuit of claim 22 wherein: (b.2b) said data processingmeans is responsive to measurement samples taken of the pulse widthmodulated output signals of the given, active channels where the timingof said measurement samples are scheduled to be in synchronism with saidreference count signal.
 24. An integrated circuit for triggering statechanges in pulsed drive signals output to a plurality of individuallycontrollable and pulse-driven operating channels, the integrated circuitcomprising: (a) a reference count generator that generates a commonreference count signal, where the common reference count signal countsthrough a sequence of successive timing values; (b) a trigger valuegenerator that generates at least one respective trigger value signalfor each pulse-driven operating channel, where the generated triggervalue signal has a respective trigger value that is to be comparedagainst the successive timing values of the common reference countsignal; and (c) comparator means for comparing the common referencecount signal against the at least one, generated and respective triggervalue signal of each pulse-driven operating channel, said comparatormeans outputting a respective comparison output signal for a givenoperating channel upon satisfaction of a predefined comparison conditionbetween compared values of the common reference count signal and the atleast one respective trigger value signal of the given operatingchannel, where the respective comparison output signal is capable ofinvoking a change in the pulsed drive signal of the given operatingchannel.
 25. The integrated circuit of claim 24 wherein the number ofindividually controllable and pulse-driven operating channels is atleast four.
 26. The integrated circuit of claim 24 wherein said triggervalue generator includes: (b.1) random access memory for storing firstdata for defining the at least one respective trigger value signal foreach pulse-driven operating channel; and (b.2) data processing means,operatively coupled to the random access memory, for producing thegenerated trigger value signals for each pulse-driven operating channelfrom said stored first data.